Universidade de Aveiro Departamento deElectr´onica, Telecomunica¸c˜oes e Inform´atica, 2018
Carlos Henrique
Cabral Medeiros
Implementation of a MIMO Transmitter for 5G
Systems
Implementa¸
c˜
ao de um Transmissor MIMO para
Sistemas 5G
Universidade de Aveiro Departamento deElectr´onica, Telecomunica¸c˜oes e Inform´atica, 2018
Carlos Henrique
Cabral Medeiros
Implementation of a MIMO Transmitter for 5G
Systems
Implementa¸
c˜
ao de um Transmissor MIMO para
Sistemas 5G
Disserta¸c˜ao apresentada `a Universidade de Aveiro para cumprimento dos requisitos necess´arios `a obten¸c˜ao do grau de Mestre em Engenharia Eletr´onica e Telecomunica¸c˜oes, realizada sob a orienta¸c˜ao cient´ıfica do Prof. Dr. Pedro Miguel da Silva Cabral, Professor Auxiliar do Departamento de Eletr´onica, Telecomunica¸c˜oes e Inform´atica da Universidade de Aveiro e sob a co-orienta¸c˜ao cient´ıfica do Dr. Luis C´otimos Nunes investigador no Instituto de Telecomunica¸c˜oes.
o j´uri / the jury
presidente / president Prof. Dr. Ant´onio Manuel Duarte Nogueira
Professor Auxiliar da Universidade de Aveiro.
vogais / examiners committee Prof. Dr. Pedro Renato Tavares de Pinho
Professor Adjunto do Instituto Superior de Engenharia de Lisboa (arguente).
Prof. Dr. Pedro Miguel da Silva Cabral
agradecimentos / acknowledgments
Agrade¸co aos meus pais e av´o, pelo esfor¸co e sacrif´ıcios que fizeram para eu concretizar o que sempre quis. O vosso amor, motiva¸c˜ao e apoio, mesmo estando longe, foi fundamental. Sem vocˆes certamente n˜ao estaria a atingir esta meta.
Ao meu orientador Prof.Pedro Cabral, um grande obrigado pelo conheci-mento transmitido, disponibilidade e, principalmente, pelo grau de exigˆencia e autonomia que me impˆos. Valores que ser˜ao sem d´uvida uma mais-valia para o meu futuro.
Agrade¸co tamb´em ao Paulo Gon¸calves pelo apoio t´ecnico, ao Filipe Barradas pela ajuda no laborat´orio e em especial ao Diogo Barros pela paciˆencia, con-selhos, conhecimentos partilhados e constante disponibilidade.
Ao Andr´e e ao Ruben, que foram a minha fam´ılia durantes estes anos, es-tiveram comigo nos melhores momentos e nos menos bons tamb´em, s˜ao por isso amigos que vou estimar e recordar para a vida. Obrigado por tudo e principalmente por terem ficado do meu lado.
`
A Claudia, que n˜ao sabe o qu˜ao importante foi, mas que esteve ao meu lado nas horas de maior dificuldade. As suas palavras de motiva¸c˜ao e apoio e a sua companhia sempre me fizeram sentir bem e motivado.
`
A Universidade de Aveiro, Departamento de Electr´onica, Telecomunica¸c˜oes e Inform´atica e ao Instituto de Telecomunica¸c˜oes por disponibilizarem tudo o que precisei para concluir este trabalho.
Resumo Nos ´ultimos anos, as comunica¸c˜oes m´oveis registaram um desenvolvimento significativo principalmente por serem cada vez mais um elemento funda-mental para o desenvolvimento econ´omico e liga¸c˜ao social. Os utilizadores definem actualmente requisitos muito exigentes, que obrigam a repensar os planos para a pr´oxima gera¸c˜ao de comunica¸c¸c˜oes m´oveis (5G). Come¸cam agora a ser feitas, previs˜oes das metas de desempenho e poss´ıveis tecnolo-gias para diversos cen´arios. Para conseguir responder `as expectativas dos servi¸cos disponibilizados por esta nova gera¸c˜ao de comunica¸c˜oes ´e necess´ario avan¸car no espectro de frequˆencia, mas as d´uvidas e desafios a ultrapassar na zona das ondas milim´etricas obriga a utilizar uma tecnologia interm´edia para implementar os primeiros sistemas 5G. Sub-6GHz ´e a tecnologia in-term´edia pensada para as primeiras aplica¸c˜oes, sendo indicada como a mel-hor combina¸c˜ao de cobertura e capacidade para as comunica¸c˜oes m´oveis com um espectro semelhante ao usado em Long Term Evolution (LTE)/4G esta tecnologia ´e capaz de atingir maiores larguras de banda e eficiˆencias espectrais mais elevadas. Os primeiros servi¸cos 5G ser˜ao entregues nas gamas de frequˆencia de 3.3 a 4.2GHz e 4.4 a 5GHz. A esta camada in-term´edia de frequˆencias, e devido `a capacidade de transmiss˜ao que se es-peram destes sistemas, vem associada a tecnologia Multiple Inputs Multiple Outputs (MIMO) para a configura¸c˜ao das antenas que vai permitir diminuir a latˆencia, aumentar a capacidade do canal de transmiss˜ao e eficiˆencia energ´etica. Cada antena na configura¸c˜ao MIMO tem o seu pr´oprio am-plicador de potˆencia que lhe faz chegar o sinal, estes ´ultimos dever˜ao ser tamb´em pequenos e extremamente eficientes. As altera¸c˜oes a n´ıvel dos materiais usados ´e tamb´em inevit´avel e o comportamento esperado dos amplificadores de potˆencia leva `a utiliza¸c˜ao de trans´ıstores de Gallium Ni-tride (GaN), que operam com caracter´ısticas mais favor´aveis a estas gamas de frequˆencias. Come¸cou a corrida aos sistemas 5G, este trabalho foca-se exatamente no estudo de um transmissor MIMO para os sistemas 5G Sub-6GHz. Para tal, ´e feito um estudo das principais caracter´ıticas dos amplificadores de r´adiofrequˆencia e dos princ´ıpios fundamentais de antenas. Este estudo ´e complementado com o projeto, implementa¸c˜ao e teste de um amplificador de potˆencia e antena e o teste destes elementos replicados em dois ramos transmissores que comp˜oem o transmissor MIMO. Desta forma, colocando em pr´atica os conceitos te´oricos e t´ecnicas de projeto estudadas, bem como a aquisi¸c˜ao de rotinas laboratoriais para o teste de dispositivos de r´adiofrequˆencia.
Abstract For the past years, mobile communications have recorded a significant development mainly because they are increasingly a fundamental element for economic and social development. Nowadays, the users define very demanding requirements forcing the re-planning for the next communication generation (5G). Predictions of performance goals and possible technologies for different scenarios are starting now to appear. In order to be able to respond to the expectations of the services provided by this new generation of communications, it is necessary to advance in the frequency spectrum, but the doubts and challenges to overcome in the area of milimeter waves requires the use of an intermediate technology to implement the first 5G systems. Sub-6Ghz is the intermediate frequency technology that starts to be developed, being indicated as the best combination of coverage and capacity for mobile communications with a spectrum similar to the one used in LTE/4G this technology is capable of reaching higher bandwidths and spectral efficiencies. It was expected that the first 5G services will be delivered in the frequency bands of 3.3 to 4.2GHz and 4.4 to 5GHz. This intermediate layer of frequencies, mainly due to the expected transmission capacity, introduces MIMO as the configuration of the antennas that will allow to decrease the latency, increase the capacity of the transmission channel and energy efficiency. Each antenna in the MIMO configuration is driven by a power amplifier, this device needs to be small in size and extremely efficient. Changes in the used materials are also inevitable and the expected behavior of power amplifiers lead to the use of GaN transistors that operate with more favorable characteristics at this frequency ranges. The run to 5G as started and this dissertation focuses exactely on the study of a MIMO transmitter for 5G Sub-6GHz systems. For this purpose, a study on the main characteristics of radio frequency amplifiers and fundamental principles of antennas is made. This study is complemented by the design, implementation and testing of a power amplifier and antenna and the test of these elements replicated in two transmitting branches that compose the MIMO transmitter. In this way, the studied theoretical concepts and design techniques are put into practice, as well as the acquisition of laboratory routines for the testing of radio frequency devices.
Contents
Contents i
List of Figures iii
List of Tables vii
Acronyms ix
1 Introduction 1
1.1 Motivation and context . . . 1
1.2 Objectives . . . 3
1.3 Thesis structure . . . 3
2 Power Amplifiers 5 2.1 Linearity and distortion . . . 5
2.1.1 1-tone Characterization . . . 5 2.1.2 AM-AM Characterization . . . 6 2.2 1dB Compression point . . . 7 2.3 AM-PM . . . 7 2.4 Two-Tone characterization . . . 8 2.4.1 Intermodulation Ratio . . . 8
2.4.2 3rd Order intercept point (IP3) . . . 8
2.5 Efficiency . . . 9
2.6 Stability . . . 10
2.6.1 Stability circles . . . 10
2.7 Memory effects . . . 11
2.8 Amplitude modulated signal effect on PA behavior . . . 11
3 Antenna design for MIMO applications 15 3.1 MIMO overview . . . 15
3.2 Fundamental parameters of antennas . . . 18
3.2.1 Radiation pattern . . . 18 3.2.2 Directivity . . . 18 3.2.3 Bandwidth . . . 18 3.2.4 Gain . . . 19 3.2.5 Polarization . . . 19 3.2.6 Input impedance . . . 19
3.3 MIMO antenna solutions . . . 19
3.3.1 Suitable Antennas for MIMO Systems . . . 20
4 Power Amplifier Design and Measurements 25 4.1 I-V curves . . . 25
4.2 Load Pull analysis . . . 25
4.3 Bias network . . . 27
4.4 Impedance Matching Networks . . . 28
4.4.1 Output Matching Network . . . 28
4.4.2 Input Matching Network . . . 30
4.5 Layout . . . 31
4.5.1 Considerations and physical constrains . . . 31
4.6 Simulated results . . . 32
4.7 Experimental validation . . . 35
4.7.1 Testing equipment, setup and procedure . . . 35
4.7.2 Measured results under a Continuous Wave (CW) input signal . . . . 36
5 Antenna Design and Measurements 39 5.1 Geometry and simulated results . . . 39
5.2 Measured results . . . 42
6 MIMO Transmitter Implementation 45 6.1 Power Amplifier modulated signal test . . . 45
6.2 Two output MIMO transmitter . . . 51
7 Conclusions and Future Work 57 7.1 Conclusions . . . 57
7.2 Future work . . . 58
Bibliography 59
List of Figures
1.1 Expectations to be met by 5G [2]. . . 2
1.2 5G systems frequency layers. . . 2
1.3 Example of a GAN PA from CREE [6]. . . 3
2.1 Two port network with input and output terminations. . . 6
2.2 1dB compression point [8]. . . 7
2.3 Illustration of AM-AM and AM-PM conversions in a nonlinear system driven by a signal of increasing amplitude envelope. y1(t): linear component; y3(t): third-order signal correlated distortion component; yr(t): resultant output component; and φ: resultant output phase.[8] . . . 8
2.4 Two tone characterization analysis[8]. . . 9
2.5 OFDM signal in time domain. . . 12
2.6 Efficiency performance curve. . . 13
2.7 Continuous Output Spectrum from Non-Linear System. . . 13
3.1 SISO communication system. . . 15
3.2 Single Input Multiple Outputs (SIMO) communication system. . . 16
3.3 MISO communication system. . . 16
3.4 MIMO communication system. . . 16
3.5 Capacity vs number of MIMO antennas. . . 17
3.6 Radiation Diagram. . . 18
3.7 Proposed UWB antena(left-frontside/right-backside) [21]. . . 20
3.8 Return loss of proposed antenna [21]. . . 20
3.9 Gain of proposed antenna. . . 21
3.10 Radiation patterns of the proposed antenna at different frequencies. . . 21
3.11 Geometry of the proposed antenna [23]. . . 22
3.12 Measured data and simulated results of the return loss [23]. . . 22
3.13 Simulated radiation patterns (a) at 4GHz and (b) at 6GHz [23]. . . 23
4.1 Characteristic DC curves (IDS vs VDS). . . 25
4.2 Characteristic DC curve for 28V drain bias (IDS vs VGS). . . 25
4.3 Load-Pull contours at 4.5 GHz (Blue - Power output; Red dotted - PAE). . . 26
4.4 Impedance at the radial stub. . . 27
4.5 Impedance after the λ/4 line. . . 27
4.6 Gate bias circuit. . . 28
4.7 Drain bias circuit. . . 28
4.9 Behavior of the DC blocking capacitor. . . 29
4.10 Tuned loads to the chosen impedance value. . . 30
4.11 Available Gain and Source Stability Circles (mRF1-4.2GHz, mRF2-4.5GHz, mRF3-4.8GHz). . . 30
4.12 Tuned loads to the chosen impedance value. . . 31
4.13 System Layout; yellow line - aluminum base; yellow circles - Screws; Red line - Teflon piece. . . 32
4.14 ADS simulated small signal transducer gain across bandwidth as a function of power output. . . 32
4.15 Momentum simulated small signal transducer gain across bandwidth as a func-tion of power output. . . 32
4.16 Advanced Design System (ADS) simulated transducer gain across bandwidth as a function of power output. . . 33
4.17 Momentum Simulated transducer gain across bandwidth as a function of power output. . . 33
4.18 ADS simulated drain efficiency as a function of power output. . . 33
4.19 Momentum simulated drain efficiency as a function of power output. . . 33
4.20 Left- Source stability circles, stable outside the circles;Right-Load stability cir-cles, stable outside the circles. . . 34
4.21 Left- Simulated reflection coefficients at the input (red) and output (blue) of the PA; Right- Stability measure and Rollett stability factor. . . 34
4.22 Wideband PA photography. . . 35
4.23 Testing setup block diagram. . . 35
4.24 Testing bench equipment picture. Legend: 1- Power supply; 2- Driver; 3- PA and Heatsink; 4- Attenuator; 5- Power Meter. . . 36
4.25 Measured transducer gain across bandwidth as a function of power output for a CW input signal. . . 36
4.26 Comparison between the Momentum simulation and the measured small signal gain for a CW input signal. . . 37
4.27 Comparison between the Momentum simulation and the measured efficiency at the 1dB compression point for a CW input signal. . . 37
4.28 Measured drain efficiency as a function of power output for a CW input signal. 38 4.29 Comparison between the Momentum simulation and the measured output power at the 1dB compression point for a CW input signal. . . 38
5.1 Antenna design (a)top view (b) bottom view. . . 39
5.2 Simulated antenna return loss. . . 40
5.3 Radiation patterns (Degree vs dBi). . . 41
5.4 3D radiation pattern. . . 41
5.5 Simulated and Measured antenna return loss.[23] . . . 42
5.6 Printed antenna. . . 43
5.7 Measured radiation patterns in the H ans E plane at 3 reference frequencies . 43 6.1 Block diagram of the test setup . . . 46
6.2 Input Signal Power Histogram. . . 46
6.3 Spectrum of 10MHz bandwidth signal at: (a)4.2GHz (b)4.5GHz (c)4.8GHz. . 48 iv
6.4 CW reasponde and AM-AM response of 10MHz bandwidth signal at: (a) 4.2GHz (b) 4.5GHz (c) 4.8GHz. . . 49 6.5 AM-PM response of 10MHz bandwidth signal at: (a) 4.2GHz (b) 4.5GHz (c)
4.8GHz. . . 50 6.6 Comparison between the two PAs drain current. . . 51 6.7 Block diagram of the test setup. . . 52 6.8 Laboratory setup. Legend: 1- AWG; 2-Drivers; 3- Power Amplifiers; 4- Heatsink;
5- Couplers; 6- FSQ; 7- Power Supply. . . 52 6.9 Laboratory setup: (a) Antenna support dimensions. (b) 1- Antenas; 2-
An-tenna support. (c) 1- AnAn-tenna Support; 2- Absorbent foam wall. . . 53 6.10 Mean efficiency values for each spacing between the antennas, varying the signal
phase. . . 54 6.11 Values of S12 between the antennas. . . 54 6.12 Comparison between the PA behavior for two signals with different phase, 0
List of Tables
4.1 PAE and power delivered at 1 dB compression point to the load impedance inside the 4.2-4.8GHz band considering 11.555-j5.103(Ω) the load impedance. 26 4.2 Simulated drain efficiency at the 1 dB compression point. . . 33 5.1 Antenna parameters. . . 40 5.2 Antenna max Gain values. . . 42 6.1 Impact of a OFDM modulated signal on 3 frequencies of the PA bandwidth. . 47
Acronyms
ACPR Adjacent Channel Power Ratio ADS Advanced Design System AM Amplitude Modulation CNC Computer Numeric Control CST Computer Simulation Technology CW Continuous Wave
DC Direct Current DUT Device Under Test GaN Gallium Nitride
GSM Global System for Mobile Communications HPBW Half Power Beam Width
IMD Intermodulation Distortion IMD Intermodulation Ratio IOT Internet of Things
LDMOS Laterally Diffused Metal-Oxide Semiconductor LTE Long Term Evolution
MIMO Multiple Inputs Multiple Outputs MISO Multiple Inputs Single Output
OFDM Orthogonal Frequency Division Multiplexing OMN Output Matching Network
PA Power Amplifier PAE Power Added Efficiency PAPR Peak-to-Average Power Ratio PCB Printed Circuit Board PM Phase Modulation PVC Polyvinyl chloride RF Radio Frequency
SIMO Single Input Multiple Outputs SISO Single Input Single Output SMA SubMiniature Version A SNR Signal to Noise Ratio VNA Vector Network Analyzer VSWR Voltage Standing Wave Ratio
Chapter 1
Introduction
The mobile communication systems have accompanied the everyday life of the world and, in the last decades, they have grown enormously. Playing a fundamental role on the connection of societies and development of their economies, this ability of communication in distance is the evolution enhancement for many branches of knowledge. For these reasons the study of these systems and the design of new ones with better characteristics are a priority.
1.1
Motivation and context
In every communication system the main concern is the energetic efficiency, this figure of merit is used to evaluate different implementations and designs as well as other system characteristics like power, linearity and bandwidth. All these requirements are needed to achieve a good coverage area, battery duration and number of available services.
Until now the known and developed generation of communication systems, 4G (fourth generation), was capable of deliver wide area coverage and good throughput while maximizing spectral efficiency. However, the increasing use of technology is leading to a fast grow of the Internet of Things (IOT) concept. By 2020, it is expected that there will be more than twenty billion devices connected, from cellphones, cars, smart houses to drones. All these devices need to share information with high data rate and maintain connectivity. 4G will not be able to ensure such capacity.
Taking in account these expectations and needs of future communication systems, the research works around the fifth generation (5G) begins. This new generation of communi-cation systems emerges not only as the foundation for advanced communicommuni-cation services but also as the infrastructure that supports socio-economic development and industrial digital transformation of the future [1].
5G should be able to manage the amount of resources available to each user depending on the devices and the services needed. Support high data traffic, around 10000 times the 4G capacity, peak rates in the range of 10 Gbps and also reduce latency by increasing the data processing are some of the expectations to be met by 5G, Figure 1.1.
Figure 1.1: Expectations to be met by 5G [2].
Future plans for 5G systems are being designed to provide to their users hundreds of MHz of bandwidth. This feature raises the need of changing the central operation frequency to the millimeter waves band (20-100GHz). However, there is still a high degree of uncertainty and challenges to overcome in this frequency zone. Therefore, some middle step technologies are being developed as backbones for the future 5G systems [3].
When defining their public position facing the future of 5G in [1], multinational com-pany of networks equipments and telecommunications, Huawei, defines three frequency layers depending on three different 5G scenarios, Figure 1.2.
In these three frequency layers, the sub-6GHz layer seems to be the best option to a middle step technology until the high frequency plans and technologies are fully developed. Sub-6GHz have a similar spectrum to the one currently used in 4G/LTE, with wider bandwidths and higher spectral efficiencies, it gives significant performance increase. For the operators the business model is already well understood and their expectation is to deploy 5G in 3.3 to 4.2GHz and 4.4 to 5GHz frequency ranges [3, 4].
Figure 1.2: 5G systems frequency layers.
In order to improve network speeds, each generation of wireless technology has used its own antenna technology. Moving forward in the frequency spectrum led to the confirmation
of the potencial that some technologies can have in the design of this new systems. MIMO is one of them. 3G used single-user MIMO, transmitting multiple data streams from the base station to a single user. Multi-user MIMO, is the dominant technology in 4G systems, it assigns different data streams to different users, increasing the capacity and performance over 3G. To achieve such high capacity, 5G systems introduce the massive MIMO. This antenna configuration, will allow to prevent transmission in undesired directions, decrease latency, increase spectral efficiency and high reliability with great energy efficiency [3].
Sub-6GHz massive MIMO is a evolving technology to deliver the first 5G services. As expected, do not come without challenges. Sub-6GHz massive MIMO will demand small, highly efficient, cost-effective PAs that can be used in the antenna arrays, since each antenna is driven by its own PA. These expected performance is leading to a change in the base stations that are switching from Laterally Diffused Metal-Oxide Semiconductor (LDMOS) to GaN PAs, Figure 1.3, since it offers many advantages at these sub-6-GHz systems [3]: GaN performs well at 3.5GHz and above, it has a high breakdown voltage, higher output impedance and lower parasitic capacitance, these properties allow to achieve high output power, wider bandwidths and higher efficiency [5]. Another important characteristic is that it can operate with higher temperatures, allowing to reduce the heatsink size.
Figure 1.3: Example of a GAN PA from CREE [6].
5G sub-6-GHz seems to be a promising technology, indicating that its study and develop-ment may be an essential step on the run to 5G systems.
1.2
Objectives
This thesis main objective is to design, implement and test a MIMO transmitter for sub-6GHz 5G systems. This implies the design of a PA and antenna, which are then replicated to form a two output transmitter. Thus, acquire experience in the design and testing techniques of Radio Frequency (RF) power amplifiers allowing to understand the physical components of the amplifying and transmitting stage of a communication system.
1.3
Thesis structure
In order to attain the objectives stipulated above, this thesis has the following structure: Chapter 1 introduces the evolution of the communications systems, the design plans and challenges of the new generation of cellular communications, as well as why was chosen a
middle step frequency technology for the design process.
In chapter 2, the essentials about the RF amplifiers theory are addressed. These con-cepts are fundamental to understand the amplifying process and the more important design techniques of RF amplifiers.
Chapter 3 gives an overview on the different antenna configuration, focusing on MIMO. An explanation on fundamental parameters of antennas and finally an overview on the state of art for antennas used in MIMO.
Chapter 4 and 5 are a design guide, simulation results and experimental validation of the PA and antenna, respectively.
The mounting process of one MIMO branch as well as the two by two MIMO transmitter and the experimental validation are presented in chapter 6.
Finally, in chapter 7 the conclusions are made facing the establish objectives and possible future work.
Chapter 2
Power Amplifiers
A PA has the objective of increasing the signal power that carries information in the communication systems. These signals have to be sufficiently high to overcome the adversities in the channel of propagation.
This signal manipulation is normally done before the propagation, being the PA one of the final components of a communication system.
In order to evaluate the performance of a PA, it can be defined different figures of merit independent from the type of architecture used when designing a PA. These are very im-portant to quantitatively evaluate the performance of a PA and the direct comparison with others [7].
In this chapter, to comprehend and design this type of circuits, the main figures of merit of a PA are going to be addressed.
2.1
Linearity and distortion
Ideally, a linear PA would have an output power proportional to the power at the input, by a constant factor of gain, independent from the excitation power. It is impossible to implement such device in practice, since it would be always limited in power by maximum value that the power supply is capable of provide. Thus, all PAs can be linear, if they operate below this limit, but the non-linear response starts to become more pronounced as the power level approaches it. At this point, the wave forms starts to saturate, the increase of output power becomes smaller, consequently it is observed a compression on gain and signal distortion.
A study on this non-linear response is essential to help the PA implementation process, this way it can be designed to attain the maximum power it is capable of deliver.
Linearity also has a fundamental role in the preservation of the electromagnetic spectrum assuring the device does not deliver power outside the frequency bandwidth where it is ex-pected to operate. Nowadays, the spectrum is very occupied with the increase of application that need space on it, becoming essential to avoid the degradation of this spectrum due to the non-linearity of the PA.
2.1.1 1-tone Characterization
A PA can easily be characterized by applying at its output a signal with a single carrier and determine experimentally some important merit figures as gain, efficiency and the analyze of
the 1dB compression point (P1dB). This type of excitation is called Continuous Wave (CW), since the RF signal envelope is constant.
This characterization is ideal to understand the behavior of the device in applications were the input signal does not have modulation in amplitude, for example the Global System for Mobile Communications (GSM).
2.1.2 AM-AM Characterization
One of the most important merit figures is gain, the relation between output and input power. Beyond indicating the power added to the input signal by the amplifier, fed by an external supply, also indicates the distortion added to the input signal in function of output power.
This nonlinear distortion on the PA can also be called Amplitude Modulation (AM)-AM conversion. It represents the modulation in amplitude of the output signal in function of the amplitude of the input signal. A lot of amplitude changes may appear in the RF path, due to temperature and power supply variations.
Figure 2.1: Two port network with input and output terminations.
Figure 2.1 illustrates a generic PA schematic, represented by the S parameters and load terminations. From the figure, there are different ways to define input power for gain calcu-lation. Therefore, three possible definitions:
• Available power gain: defined based on the power available at the output of the amplifier (PAV N) and the power available from the source (PAV S), it assumes max power transfer between the amplifier and the load. Not reasonable for practical use, as there are no ideal components.
GA= PAV N PAV S
(2.1) • Operating power gain: defined by power delivered to the load (PL) and the power at the input of the quadrupole (PIN). It assumes max power transfer between the source and the amplifier’s input.
GP = PL PIN
(2.2) • Transducer power gain: defined based on the power delivered to the load (PL) and the power available from the source (PAV S). Takes in account every possible loss and does
not assume any condition on power transfer, therefore the most used definition. GT = PL PAV S (2.3)
2.2
1dB Compression point
When operating in the linear region, the relation between output and input power is directly proportional. However, out of this region a direct relation cannot be establish.
In the PA, for small signal the behavior is almost linearly, when the input is increased. This device, like the most of nonlinear devices reaches saturation and losses become apparent. A point is reached when the output power of the device is not increased by the same amount as the input power, this point defines the compression point of power at the output. 1 dB point is identified as the output power point where the gain has dropped 1 dB. Is then used to identify the limit of the linear region.
Load pull simulations are frequently used to identify the load value that allow to attain the max value for P1dB. This value is understood as optimum when the objective is to extract the max power from the device with a reasonable distortion level.
In Figure 2.2 the dotted line is the response of the device if it was linear. The solid line represents the output power. P(1dB) is defined as the point from where the two lines diverge by 1dB.
Figure 2.2: 1dB compression point [8].
2.3
AM-PM
AM-Phase Modulation (PM) represents the phase distortion of the output signal in func-tion of the amplitude of the input signal.
Behaviorally, this type of distortion is different depending on the type of active device and it has origin in the non-linear elements which constitute it, for example the capacities CGS,CDS and CGD. The distortion components generated may not add up in phase with the linear component, introducing a phase change in the resulting vector that varies with the amplitude, Figure 2.3 [8, 9].
AM-PM conversion grows as the output power approaches the P1dB once the distortion components become increasingly significant as the PA saturates.
A detailed analysis on the origin of this type of distortion, as well as the AM-AM is made in [9] for the LDMOS technology and in [10], additionally, for the GaN technology.
Figure 2.3: Illustration of AM-AM and AM-PM conversions in a nonlinear system driven by a signal of increasing amplitude envelope. y1(t): linear component; y3(t): third-order signal correlated distortion component; yr(t): resultant output component; and φ: resultant output phase.[8]
2.4
Two-Tone characterization
Feeding the PA with a signal constituted by two or more tones, this is, the sum of two or more sinusoids at close frequencies f1 and f2, thus, two signals of frequency f1 and f2, led to the appearance of distortion components. If the signals are close together in frequency, some of the sum and difference frequencies called intermodulation products produced can occur within the bandwidth of the amplifier. These cannot be filtered out, so they will ultimately become interfering signals to the main signals to be amplified, Intermodulation Distortion (IMD). When the signal has only one frequency but is modulated in amplitude the same happens[11]. Figure 2.4a shows two signals f1 and f2 occurring within the amplifier bandwidth. With distortion, new signals f1 – f2 and f1 + f2 are produced. They can usually be filtered out. However, these signals will also mix with the second, third, and higher harmonics to pro-duce a wide range of potentially interfering signals with the amplifier pass band. The most troublesome are the third-order products, which are 2f1 ± f2 and 2f2 ± f1 .
2.4.1 Intermodulation Ratio
Intermodulation Ratio (IMD) is the ratio between the fundamental tone power and the distortion tone close to the useful band. For systems were the distortion in the lateral bands is asymmetric, is necessary to define IMD for both. It can be calculated by the expression,
IM R = Pf und PIM D = P (w1) P (2w1 − w2) = P (w2) P (2w2 − w1) (2.4)
In weak signal zone the output signal rises 1 dB for each 1dB increase in the input power, however the distortion power increases 3 dB. Therefore, the IMD decreases with the increase of Pin.
2.4.2 3rd Order intercept point (IP3)
A figure of merit that characterize the IMD through the interception point between the power at the fundamental and the IMD, extrapolated for the strong signal regime from the weak signal regime.
The 3rd order intercept point, Figure 2.4b, is a theoretical point where the power of the third order distortion signal would equal the power of the fundamental signal. This point is
never reached, as the amplifier will saturate before this condition can occur. Nevertheless, it is a good indicator of amplifier linearity degradation [11].
(a) IMD products.
(b) 3rd order intercept point
Figure 2.4: Two tone characterization analysis[8].
2.5
Efficiency
Efficiency is a measure of how well a device converts one energy source to another. Is one of the most important measures of a PA, as it determines the capability of the device transforming Direct Current (DC) power into RF power. What doesn’t get converted to power goes into heat. There are three ways of defining this merit factor:
• Drain Efficiency: is a measure of how much DC power(PDC) is converted to RF power (Pout). The problem with using this measurement as a benchmark is that it doesn’t consider the incident RF power that goes into a device.
η = Pout PDC
(2.5)
• Power Added Efficiency (PAE): subtracts the input power from the output power, considering only the fraction of the output power that was obtained through the DC supply. The most important efficiency for RF devices.
P AE = Pout− Pin PDC
(2.6)
• Total efficiency: Determines the ratio of output power to the total power supply. With this definition it is possible to characterize the efficiency of a multiple stage amplifier where there are several devices consuming power.
ηtotal=
PRF out PDC+PNn=0Pn
2.6
Stability
When projecting a PA is necessary to assure in the end the device is stable since it can present input and output negative impedances. If the PA becomes unstable it amplifies the input signal with increasing gain until saturation and may cause damage if the limit of power dissipation is surpassed. The stability of an amplifier, or its resistance to oscillate, is a very important consideration in a design and can be determined from the S parameters, the matching networks, and the terminations. In [13] it is proved that the input and output reflection coefficients can be defined from following equations,
ΓIN = S11+ S12S21ΓL 1 − S22ΓL (2.8) ΓOU T = S22+ S12S21ΓS 1 − S11ΓS (2.9) A device can be characterized as unconditionally unstable if the following conditions are met,
|ΓOU T| < 1 ∧ |ΓIN| < 1 (2.10) In a passive device the input and output impedances are always positive guaranteeing the absence of oscillations. On the other hand, if is an active device then S12 6= 0 and S22 can be sufficient high to make |ΓOU T| > 1 e |ΓIN| > 1. If the active device is unilateral, thus S12 = 0, the reflection coefficients still depend on S11 and S22 whose absolute value might not be inferior to one and the device is potentially unstable.
The stability of a device can be analyzed applying the Rollet stability factor (K): (
K = 1−|S11|2−|S22|2+|∆|2
2|S12S21|
|∆| = |S11S22− S12S21|
(2.11) From the previous equations if K > 1 and ∆ < 1 the device is unconditionally stable. If K > 1 the PA is conditionally stable [13].
2.6.1 Stability circles
Stability can also be analyzed alongside with Rollet Factor, to identify , in the Smith Chart, which terminations assure stability. From the expressions in 2.12, the limit of stability are verified as,
|ΓOU T| = 1 ∧ |ΓIN| = 1 (2.12)
this conditions form a circles whose radius and center can be determined by rL= S12S21 |S22|2− |∆|2 (2.13) and CL= (S22− ∆S11∗ )∗ |S22|2− |∆|2 (2.14) where rL nd CL are the radius and center of the circles mapped in the ΓL plane and finally,
rS = S12S21 |S11|2− |∆|2 (2.15) 10
and
CS=
(S11− ∆S22∗ )∗
|S11|2− |∆|2 (2.16)
where rS nd CS are the radius and center of the circles mapped in the ΓS plane.
When plotted in a Smith Chart, the circles delimit two areas, one stable and the other potentially stable. The last step on the stability analysis is to determine which one is the stable region. The ΓIN and ΓOU T can be calculated for the center of the circle. If the stability condition is verified the stable zone is the interior of the circle, otherwise is the outside. But it is sufficient to test the stability condition in the center of the Smith Chart, where ΓS = 0 and ΓL = 0. Then ΓIN = |S11| and ΓOU T = |S22|. This way the test of stability is based on the absolute value analysis of the input and output reflection coefficients, if the stability condition is verified then the stable region is the inside of the circle otherwise is the outside.
2.7
Memory effects
A PA will show some deviation from its static characteristics. Such deviation have become known as ”memory effects”. Effects on a timescale that can be associated with the envelope or information rate of the signal, and any variations in the PA performance or behavior on this timescale leads to distortion of the signal. Memory effects are an additional source of nonlinear behavior that is typically not accounted for the PA models. The origin of memory effects can be: Inherent to the active device itself as Thermal Effects, Trapping and imposed by external circuitry as bias networks [14].
• Thermal effects: As the amplitude of the signal changes in time so the input energy, leading the transistor do heat up and cool down. This rates of heating and cooling are dependent on the semiconductor material of the transistor. As a result the instantaneous gain of the PA changes in time, generating a complex gain behavior that is dependent not only on the signal value at a given instant, but also on the recent history of the signal [15].
• Trapping: The capture and emission of electron in the transistor channel, causes changes in the current flowing trough the device that is again depending not only on the instan-taneous voltage, but also on the history voltage of the device.
• Bias circuits: The Printed Circuit Board (PCB) traces, the capacitors found on the DC bias and supply circuits on the PA contribute to inductive and capacitive energy storage that cause the memory effects.
These memory effects are visible in the AM-AM and AM-PM response, when a spreading of the measured values happen around the static values.
2.8
Amplitude modulated signal effect on PA behavior
Further ahead, the designed PA will operate under a Orthogonal Frequency Division Multiplexing (OFDM) signal, so it is important to undestand which behavior should be expected.
Figure 2.5: OFDM signal in time domain.
OFDM is a Multi-Carrier Modulation technique in which a single high rate data-stream is divided into multiple low rate data-streams and is modulated using subcarriers which are orthogonal to each other. It is an advanced modulation technique which is suitable for high-speed data transmission due to several advantageous features : High spectral efficiency, immunity to impulse interferences or robustness to channel fading
In spite of these benefits there are some obstacles in using OFDM, such as intercarrier interference or the main problem for the PA performance which is the Peak-to-Average Power Ratio (PAPR).
The PAPR is defined as the relation between the signal maximum power and its average power. Signals as OFDM with high PAPR values present most of the time low and medium power levels and only reaches the maximum efficiency value (maximum power) occasionally, Figure 2.5.
When signals with this characteristic are used in a PA, this device should be operated in a very large linear region. Otherwise, the signal peaks get into non-linear region of the power amplifier causing signal distortion. Thus, the power amplifiers should be operated with large power back-offs, causing very inefficient amplification and lower efficiency, Figure 2.6. These large peaks can also cause saturation in power amplifiers, leading to intermodulation products among the subcarriers [16].
Figure 2.6: Efficiency performance curve.
When working with modulated signal, it is also important to analyze the Adjacent Chan-nel Power Ratio (ACPR). The ACPR is the equivalent to the IMD for continuous signal in the frequency domain. The ACPR integrates the power along all of the adjacent channels and ratios in relation to the main channel power [12]. The adjacent channel may be considered anywhere, usually the worst case in taken in account, Figure 2.7. The ACPR can be calcu-lated, equation 2.17 by integrating the adjacent channels to infinity, in a limited bandwidth or even subdividing the adjacent channels and calculating an ACPR for each [12].
ACP R = R BwSadjacentdw Rwlow whighSmaindw (2.17)
Chapter 3
Antenna design for MIMO
applications
An antenna is a passive element from radio systems used to receive or radiate electromag-netic waves. In wireless systems it is one of the most important elements, crucial to determine the system performance.[17]
Thus, it is important to find out the proper design for the antenna to satisfy the require-ments of the final system.
Firstly, different antenna configurations that led to MIMO are going to be addressed. Then, a review on the most important antenna properties that must be confirmed to ensure a good performance. Finally, a review on already designed antennas for MIMO systems.
3.1
MIMO overview
As seen before, the expectation of data rates and efficiency delivered by future communi-cations systems are becoming higher, and the capacity must be increased, since it determines the quality of the communication system.
Different models of antenna configuration can be defined when referring to the number of input and output antennas on each side of a communication channel.
The simplest configuration is a Single Input Single Output (SISO) system, in which there is a single transmitting antenna at the source and a single receiving antenna at the destination, Figure 3.1. Systems of this type are commonly used in Bluetooth, Wi-Fi, TV. The simplicity of the system leads to multipath effects since the send wave interacts with the surroundings arriving to the destination from many paths. Consequently, this effect causes fading, less data speed, and errors in the data received [18] .
Figure 3.1: SISO communication system.
SIMO, where the receiver is equipped with multiple antennas, Figure 3.2, ensuring di-versity to combat fading effects. This type of configuration implies didi-versity schemes in the receiver, to improve Signal to Noise Ratio (SNR) such as Combining Selection (system selects
the strongest signal received), Equal Gain Combining (all received signals are combined re-sulting in a higher SNR and Maximum Ratio Combining (the strongest signals are summed and combined). The processing may be limited in size, cost and battery when used in mobile devices.[18]
Figure 3.2: SIMO communication system.
Multiple Inputs Single Output (MISO) or the multiple input and single output is a wireless system with multiple transmitting antennas and a single antenna at the destination, Figure 3.3. Like SIMO, the multipath effects are reduced. The redundancy and processing are moved from the receiver to the transmitter.
Figure 3.3: MISO communication system.
MIMO systems or the multiple input and multiple output systems, use multiple antennas at the transmitter and receiver, Figure 3.4, being very powerful in performance-enhancing of systems capabilities presenting channel robustness and throughput.
Figure 3.4: MIMO communication system.
MIMO technology constitutes a breakthrough in wireless communication system design. The technology offers several benefits that help meet the challenges placed by both the im-pairments in the wireless channel as well as resource constraints [19]:
• Array gain
Combining the signals at the receiver results in a increase of the SNR. This coherent combining may be realized by spatial processing at the receive antenna array and/or spatial pre-processing at the transmit antenna array. Array gain improves resistance to noise. Thus, better coverage and range of a wireless network.
• Spatial diversity gain
When providing multiple copies of the transmitted signal in space, frequency or time it is possible to reduce the fading effects. This number of independent copies increase the probability that at least one of the copies does not experience a deep fade effect, improving the quality of reception. The number of independent possible links is given by the special diversity order NRxNT (NT, number of transmit antennas and NR, number of receiving antennas) [19].
• Spatial multiplexing gain
With spatial multiplexing, independent data streams are transmitted within the band-width of operation, then the receiver separates de data streams. At the end, each data stream experiences at least the same channel quality that would be experienced by a SISO system, enhancing the capacity by a multiplicative factor equal to the number of streams.
In general, the number of data streams that can be reliably supported by a MIMO channel equals the minimum of the number of transmit antennas and the number of receive antennas, The spatial multiplexing gain increases the capacity of a wireless network, Figure 3.5 [19].
3.2
Fundamental parameters of antennas
3.2.1 Radiation pattern
Defined as a graphical representation of the radiation properties of the antenna as a function of space coordinates.
An isotropic radiator has a radiation pattern with equal radiation in all directions. If the radiation is constant but with a maximum in a certain plain, the antenna is omnidirectional. When it can be identified a main lobe of radiation then the antenna is directional.
A lobe is a portion of the radiation pattern limited by region of weaker radiation intensity. The main lobe defines the direction of maximum radiation, the other lobes are secondary. The characterization of the main lobe is used to compare antennas, through the Half Power Beam Width (HPBW), Figure 3.6, the angle subtended by the half power-point of the main lobe [17].
Figure 3.6: Radiation Diagram.
3.2.2 Directivity
Directivity is defined as the ratio of the radiation intensity in a given direction from the antenna to the radiation intensity averaged over all directions.
Simply stated and from the mathematical equation 3.1, directivity of a non-isotropic source is equal to the ratio of its radiation intensity in a given direction over that, of an isotropic source.
D(θ, φ) = 4πU (θ, φ) Prad
(3.1) The directivity increases when the radiation angle is smaller.
3.2.3 Bandwidth
Bandwidth is defined as the range over which the antenna present good performance taking in account the requirements. It was defined for this dissertation, the bandwidth would be the frequency range where the return loss would be below -10 dB and the Voltage Standing Wave
Ratio (VSWR) lower than 2. The radiation pattern of an antenna may change dramatically outside its specified operating bandwidth.
3.2.4 Gain
Gain of an antenna is defined as the ratio of the intensity, in a given direction, to the radiation intensity that would be obtained if the power accepted by the antenna was radiated isotropically. [17] This reference antenna is in most cases a lossless source.
G(θ, φ) = 4πU (θ, φ) Pin
(3.2) It can be written that Prad= ηPin, meaning gain is related with directivity and efficiency.
3.2.5 Polarization
In [17] polarization of a wave is defined as the curve traced by the end point of the vector representing the instantaneous electric field observed along the direction of propagation. The polarization of an antenna can be defined as linear, circular and elliptical.
If the electric filed vector is always oriented along the same straight line at every instant time, the time-harmonic wave is linearly polarized.
In the circular polarization the electric filed presents two orthogonal linear components with the same magnitude, with a phase difference of odd multiples of 90. This type of polarization can be classified in other two subtypes, clockwise or counter clockwise depending in the rotation direction of the electric filed.
The elliptical polarization appears when the two orthogonal components mentioned above have different magnitudes.
It is important that in an antenna system the transmitter and receiver antenna have the same type of polarization, or the system will have losses from polarization.[17]
3.2.6 Input impedance
Defined as the impedance presented by an antenna at its terminals, it varies with the frequency, so it is only possible to match this impedance value in a certain frequency band.
The value of this impedance is normally difficult to obtain trough mathematical equations, so is obtained by simulation. The input impedance has a real and an imaginary part.
Za= Ra+ jXa (3.3)
Ra, represents the radiated power from the antenna and jXa the stored power from the nearby antenna field.
3.3
MIMO antenna solutions
Plans for the future mobile communications systems are that, the base station and even the user’s terminal will be equipped with multi-antenna systems. This specification implies that the antennas need to be smaller and cheaper.
Working in a frequency band around 4.5 GHz and a bandwidth of 600 MHz, printed anten-nas appear to be the best choice, due to its low-cost, easy fabrication and small dimensions.
Printed Antennas, in its basic form, are a conduction patch of planar or non-planar ge-ometry on one side of a dielectric substrate with a ground plane in the opposite side. There are three main types of printed antennas: traveling wave, patch and slot antennas. [20]
3.3.1 Suitable Antennas for MIMO Systems
Several investigations on wideband printed patch antennas suitable for MIMO applications have been reported.
A simple patch antenna does not have the bandwidth needed for a communication system, since usually this antennas have a narrow bandwidth and are designed for a single frequency value. Increase the bandwidth can be done by cutting the ground plane, studies as [21] [22], proved this technique can in fact increase the bandwidth of the antenna. As consequence, the radiation patterns turn into omnidirectional patterns and the gain in the direction of the communication decrease.
In [21] is proposed an antenna, which consists of a square patch, a partial ground plane and a single rectangular slot on the ground plane. Three techniques are used to tune the return loss and bandwidth over a wide range of frequency: the use of square radiating patch, a partial ground plane and a single slot on the ground plane, Figure 3.7.
Figure 3.7: Proposed UWB antena(left-frontside/right-backside) [21].
The bottom of the square patch is connected by a microstrip line, which is fed by a 50 ω coaxial probe from the side of the antenna. The microstrip line was etched on the same side of the substrate as the radiator.
Figure 3.8: Return loss of proposed antenna [21].
Figure 3.8 shows the impedance bandwidth of 12.49GHz. This antenna has a very low gain, Figure 3.9, since this characteristic is affected by the size of the ground plane.
Figure 3.9: Gain of proposed antenna.
As expected the partial ground plane led to bidirectional radiation patterns with main beam variating between 0 and 180 degrees .
Figure 3.10: Radiation patterns of the proposed antenna at different frequencies.
In [23], a proximity feed technique is used, in which a 50Ω microstrip line is placed in one side of the substrate of the antenna facing the patch that is placed in the other side of the substrate. This patch is non-concentrically placed inside the ground plane slot, Figure 3.11.
Figure 3.11: Geometry of the proposed antenna [23].
The antenna was designed to achieve impedance bandwidth, gain and stable radiation patterns across the whole desired band. A design made in FR4 substrate with the relative permittivity of 4.7, a thickness of 1.5mm and a size of 30(L) × 30(W )mm2.
Figure 3.12: Measured data and simulated results of the return loss [23].
From Figure 3.8, it is possible to verify that the antennas has indeed a wideband response. The measured results presented a frequency band of 4.5GHz. The radiation patterns are also investigated and this prototype antenna is characterized by omnidirectional patterns in E-plane, while it is a quasi-omnidirectional pattern in H-E-plane, Figure 3.13. Achieving a gain of 4dB across the bandwidth [23].
Figure 3.13: Simulated radiation patterns (a) at 4GHz and (b) at 6GHz [23].
This geometry will be used hereafter as a reference for the design process of an antenna that meets the requirements of this dissertation.
Chapter 4
Power Amplifier Design and
Measurements
In this chapter a class B power amplifier is designed for a center frequency of 4.5GHz with a 600MHz bandwidth. The available active device was CGH40010F from CREE Inc, and the substrate was Isola Astra. The design was made using the simulation software ADS from Keysight Technologies.
4.1
I-V curves
Biased in order to achieve a class B operation PA, as a compromise between efficiency, power delivered to the load and linearity.
From Figures 4.1 and 4.2, the polarization point can be chosen. The manufacturer recom-mended 28V drain voltage, to maximize voltage excursion without reaching the breakdown voltage, so for a class B of operation a -3.1V gate voltage will be used in the design.
Figure 4.1: Characteristic DC curves (IDS vs VDS).
Figure 4.2: Characteristic DC curve for 28V drain bias (IDS vs VGS).
4.2
Load Pull analysis
Load Pull refers to presenting a priori impedance to a Device Under Test (DUT) in a controlled way, in order to extract the optimal performance from the DUT. Optimal
performance is found by varying terminal impedances along with frequency. Thus, load pull systems enable rapid, accurate and reliable determination of PA’s performance parameters.
The ADS 1-tone Load Pull for 1 dB Gain compression point simulation template is used in order to understand what are the optimum loads throughout the bandwidth to achieve maximum PAE and enough power delivered to the load.
The large signal model provided by Cree is used under CW excitation, producing Fig-ure 4.3, efficiency and output power contours. The center of each contour represents the impedance that maximizes PAE (red dotted contours) and output power (blue contours). Each counter characterizes the set of impedances, that will have the same PAE or output power, when presented to the device.
In order to obtain a good compromise between PAE and output power delivery, the impedance selected should be favorable for both group of contours. For the center frequency, the optimum load was 11.555-j5.103 (Ω), this was also the load considered for the remaining frequencies producing the PAE and power delivered presented in Table 4.1.
Figure 4.3: Load-Pull contours at 4.5 GHz (Blue - Power output; Red dotted - PAE).
Freq (GHz) PAE (%) Power delivered (dBm)
4.2 68 42.28 4.3 68.81 42.15 4.4 69.36 41.99 4.5 69.47 41.79 4.6 69.27 41.55 4.7 68.88 41.30 4.8 68.34 41.03
Table 4.1: PAE and power delivered at 1 dB compression point to the load impedance inside the 4.2-4.8GHz band considering 11.555-j5.103(Ω) the load impedance.
4.3
Bias network
One of the most important aspects to attend when designing a PA is to feed the transistor with the appropriate bias point.
The objective of the DC bias is while allowing the DC to pass to the gate and drain of the transistor, present a high impedance at the fundamental and a short circuit to the second harmonic. Furthermore, since PA’s are known for having oscillatory behavior, which can damage the device, a biasing circuit plays a major role preventing oscillation by eliminating negative resistance seen into either the input or output port.
The harmonic filtering can be done using RF capacitors, but this can cause unwanted reflections and losses due to their parasitic resistance, inductance and capacitance.The higher the frequency, the most these parasitics are observable. Instead of using capacitors, radial stubs were used to produce the same effect.
Two radial stubs were dimensioned, to create a short circuit at the central frequency 4.5GHz, Figure 4.4, and at the 9GHz, the second harmonic. After the stubs a quarter wave-length line was added to create an open circuit at the fundamental, Figure 4.5. At the second harmonic this line has half wavelength, therefore, a short circuit at this frequency.
Since there is a need for connections T’s between the lines and stubs, every time a con-nection was added a tune operation, on the stubs and on the quarter wavelength line, was made to adjust the impedance value to the expected.
The bias circuit for the gate of the transistor follows the same procedure, but a resistor is added to increase stability. The active device shows negative resistance at the input and output bias ports, and this may cause oscillations at low frequencies. Adding a resistor in the input cancels the negative resistance at this port and reduces the magnitude of the negative resistance at the output. At higher frequencies the bias present a high impedance, so the resistor has no effect. Both networks are presented in Figures 4.6 and 4.7.
Above the stubs towards the power supply, where the power cables are welded, 3 capacitors (47µF, 470 nF and 47 nF) were added to guarantee the decoupling. There is no RF signal in this section of the circuit.
Figure 4.6: Gate bias circuit. Figure 4.7: Drain bias circuit.
4.4
Impedance Matching Networks
Amplifying the signal that flows from the source to the load is the main objective of the PA, Figure 4.8. The amplifying process needs to be efficient, thus reflection of the same signal must be reduced , otherwise can destroy the quality of the signal and reduce the output power. These reflections can be avoid using matching networks, which matches the impedances at the active devices ports.
Figure 4.8: Block of complete PA.
4.4.1 Output Matching Network
As seen before with the load pull results, it is possible to achieve a good compromise between efficiency and output power. This way, being the impedance at the center frequency close to the impedances along the bandwidth, 11.555+j5.103(Ω) was the value chosen to present at the transistor drain, since it allows to achieve good drain efficiency and power
delivered in the bandwidth as seen in Table 4.1. The Output Matching Network (OMN) is designed in order to present the correct impedance throughout the 600MHz bandwidth.
In [11], it is explained that the matching circuit may be obtained from a filter circuit, following a set of equations, since the impedance transformation is independent of frequency. The first step do design the OMN is then to design a low-pass filter. From [24], a real-to-real Chebyshev low pass filter is the starting point, the prototype in lumped elements extracted is then transformed into transmission line.
Using the ADS optimizer, it is possible to start from a small order filter already with transmission lines and automatically and iteratively adjust their dimensions and increase the order when needed. This process was carried out taking into account the bias already designed.
Furthermore, to block the DC component of reaching the load, a 2.2pFRF capacitor from AT Ceramics was used (800A100), Figure 4.9. The capacitor was carefully chosen in order not to block RF, especially the frequencies inside the band of operation.
Figure 4.9: Behavior of the DC blocking capacitor.
As seen above the objective of the OMN is to tune the load to the one chosen in the Load Pull process and at the same short circuit the harmonics. Broadband optimizations were done until a small network was achieved, to prevent losses in the substrate. Figure 4.10, presents the load values for each frequency optimized for the chosen value. Since, the loads varies with frequency, it is not possible to achieve the optimum load at all frequencies, but the closer value possible. This difference will cause a change on the expected output power obtained in the Load Pull simulation.
Figure 4.10: Tuned loads to the chosen impedance value.
4.4.2 Input Matching Network
After finishing the OMN, it is necessary to measure the reflection coefficient at the gate of the transistor, already with the matching and bias network at the output.
Figure 4.11 shows the available gain circles for the bandwidth frequencies. The intersection of the circles defines the conjugate impedance value that should be presented to the transistor input to maintain a constant gain throughout the bandwidth, maximize the transconductance gain and improve PAE. The chosen impedance allow to achieve a max 13.3 dB gain.
Figure 4.11: Available Gain and Source Stability Circles (mRF1-4.2GHz, mRF2-4.5GHz, mRF3-4.8GHz).
Figure 4.12: Tuned loads to the chosen impedance value.
Figure 4.12, presents the impedances at each frequency inside the bandwidth tuned for the impedance value. Once again, it was not possible to achieve the exactly expected value, but a value that was close enough.
The design process is the same described for the OMN.
A filter, resistor and RF capacitor in parallel, was added in the input network to prevent oscillations. At DC, the capacitor is a open circuit and the resistor decreases the gain at low frequencies and at high frequencies this circuit is a parallel between two impedances where the resistor higher than the capacitor impedance, does not have effect.
4.5
Layout
4.5.1 Considerations and physical constrains
In the design process of the bias circuits and matching networks, there was the need to take in account some physical constrains, namely the components spacing and line positioning.
The transistor is hold in place using a piece of Teflon, for easy removal of the device, screwed to a aluminum board under the PA circuit board. The microstrip lines had to be carefully design not to make contact with the screws.
The final step of the design was to produce the layout of the complete system, Figure 4.13, and confirm all the measurements and spacings. An electromagnetic simulation was carried out in Momentum to confirm the previously obtained results and a comparison between this and ADS simulation lay on Simulated results section.
Figure 4.13: System Layout; yellow line aluminum base; yellow circles Screws; Red line -Teflon piece.
4.6
Simulated results
After completing the design and simulation of the PA, it is possible to conclude, according to the graphics analysis, that the Momentum and ADS simulation are very similar.
Figure 4.14 and 4.15 proves small signal transducer gain to be at around 10.5 ± 0.5dB in both simulations. A typical class B gain shape can be observed in Figure 4.16 and 4.17 .
Figure 4.14: ADS simulated small signal transducer gain across bandwidth as a func-tion of power output.
Figure 4.15: Momentum simulated small sig-nal transducer gain across bandwidth as a function of power output.
Figure 4.16: ADS simulated transducer gain across bandwidth as a function of power out-put.
Figure 4.17: Momentum Simulated trans-ducer gain across bandwidth as a function of power output.
The obtained drain efficiency values averaged 63% through the band of operation, for both the simulations which once again are very similar, as shown in Table 4.2 and Figures 4.19 and 4.18.
Figure 4.18: ADS simulated drain efficiency as a function of power output.
Figure 4.19: Momentum simulated drain effi-ciency as a function of power output.
Freq (GHz) Drain Efficiency ADS (%) Drain Efficiency Momentum (%)
4.2 65 65 4.3 64 64 4.4 62 62 4.5 58 60 4.6 63 62 4.7 65 65 4.8 63 63
Table 4.2: Simulated drain efficiency at the 1 dB compression point.
circles from Figure 4.20, this circles prove that a 50Ω source or load can be connected to the input or output without causing instability since this loads are in the stable region, outside the circles.
Furthermore the magnitude of the input and output reflection coefficients, are less than one, stability measure is positive and stability factor bigger then the unit, Figure 4.21. Thus the PA is unconditional stable.
Figure 4.20: Left- Source stability circles, stable outside the circles;Right-Load stability cir-cles, stable outside the circles.
Figure 4.21: Left- Simulated reflection coefficients at the input (red) and output (blue) of the PA; Right- Stability measure and Rollett stability factor.
Completing the stability analysis , the PA is ready to be printed and tested. The experi-mental results will be addressed in the next section.
4.7
Experimental validation
4.7.1 Testing equipment, setup and procedure
The produced wideband PA, displayed in Figure 4.22, was tested in order to confirm the simulation results. The testing bench is constituted by a signal generator (Rohde & Schwarz SMW200A), to generate an signal which is amplified by a commercial PA (Aethercomm, SSPA 0.020-6.000-10) driving the PA under test, and an output 46dB attenuator to prevent damage on the power analyzer (Agilent Technologies,N1913A). Power supplies were also used to provide power for the PAs and heatsinks. Figure 4.23 shows the block diagram of the test setup and Figure 4.24 displays the testing bench used for the measurement.
The transistor was biased with VDS = 28V and VGS = −2.65V to achieve 35mA of drain current as designed.
Figure 4.22: Wideband PA photography.
Figure 4.24: Testing bench equipment picture. Legend: 1- Power supply; 2- Driver; 3- PA and Heatsink; 4- Attenuator; 5- Power Meter.
4.7.2 Measured results under a Continuous Wave (CW) input signal
A CW signal, from 4.1GHz to 4.9GHz, 100MHz before and past the limits of the band-width, with 50MHz increments, was applied to the PA’s input with increasing power until, approximately, 3dB of gain compression at the output of the PA, that was not always achieved. The measured results are presented hereafter.
Small signal gain varied approximately 1.86dB, between 9.6dB and 11.54db inside the bandwidth. The measured gain follows the evolution of simulation gain, being higher through-out the band (Figure 4.26). The most significant difference is between 4.2GHz and 4.35GHz, around 1.5dB, from there the difference is approximately 0.5dB. The shape of the gain curves in Figure 4.25 show the PA is operating in class B, for easier analysis the showed frequencies are the tested frequencies with a 100MHz steps .
Figure 4.25: Measured transducer gain across bandwidth as a function of power output for a CW input signal.
Figure 4.26: Comparison between the Momentum simulation and the measured small signal gain for a CW input signal.
The measured efficiency values at the P1dB, Figure 4.27, are in average 6% below the simulated ones. The major deviation inside the bandwidth is 9%, registered at 4.7GHz and 4.75GHz. This discrepancy between simulated and measured results might be due to manufacturing imperfections in the OMN.
Figure 4.29, shows that the output power obtained is lower than the simulated results, which causes the already seen reduction on the drain efficiency.
Figure 4.27: Comparison between the Momentum simulation and the measured efficiency at the 1dB compression point for a CW input signal.
Figure 4.28: Measured drain efficiency as a function of power output for a CW input signal.
Figure 4.29: Comparison between the Momentum simulation and the measured output power at the 1dB compression point for a CW input signal.
Chapter 5
Antenna Design and Measurements
As seen in chapter 3, for the proposed range of frequencies, microstrip antennas can have advantages such as, a simple and light body, adjustable radiation pattern, relatively easy manufacturing process, and materials of cost.In this chapter a wideband antenna is designed to have a bandwidth (|S11| ≤ −10dB) from 4GHz to 5GHz, this range covers the PA bandwidth.
The antenna was designed and simulated in Computer Simulation Technology (CST) Microwave Studio Software.
5.1
Geometry and simulated results
The manufacturing of the wideband antenna had as reference the antenna proposed in [23], analyzed in Chapter 3. Designed and printed in a FR4 substrate of 1.6mm thickness and εr = 4.3, in one side of the antenna a slotted ground plane with a round patch inside it, proximity fed by a 50Ω line. The fed line was divided in two 100Ω line, creating a fork shaped tuning stub to help achieve broad impedance matching, Figure 5.1.
As seen before in Chapter 3 the reference antenna has an omnidirectional radiation pattern and a max gain of 4dB. With the main objective of increasing the max gain and changing the radiation patterns a reflector plan was added facing the bottom side of the antenna.
The dimensions presented in Table 5.1 are the final optimized dimensions of the antenna taking in account the reflector plan and the SubMiniature Version A (SMA) 50Ω connector. It is possible to verify the antenna has small dimensions, an important characteristic since the MIMO systems are set, as the name implies, with multiple antennas.
Parameter Dimension (mm) W 30 L 34 R1 7 R2 12.65 wf 3.27 lf 11.35 ws 0.4 ls 3.6 wd 8.13 Reflector dimensions 50x50 Table 5.1: Antenna parameters.
One of the initial and most important goals is to have a good return loss response ,|S11| ≤ −10dB, inside the frequency bandwidth. Figure 5.2, presents the return loss obtain between 3GHz and 6GHz where is possible to verify that between 3.6GHz and 5.1GHz, the return loss is less than -10dB, value from which it is possible to define impedance bandwidth, since inside it less of one-tenth of the transmitted energy is lost by reflection.
Figure 5.2: Simulated antenna return loss.
Since this is an antenna with a defined bandwidth, according to the obtained results, between between 3.6GHz and 5.1GHz, in terms of radiation patterns, 3 reference frequencies
