Development of a remote radio unit and optical access link for the ORCIP C-RAN testbed

(1)

Universidade de Aveiro Departamento deEletrónica, Telecomunicações e Informática 2019

Bruno Tavares

Brandão

Desenvolvimento de uma Unidade de Rádio

Remota e Rede Ótica de Acesso para a

Infraestrutura ORCIP C-RAN

(2)

(3)

Universidade de Aveiro Departamento deEletrónica, Telecomunicações e Informática 2019

Bruno Tavares

Brandão

Desenvolvimento de uma Unidade de Rádio

Remota e Rede Ótica de Acesso para a

Infraestrutura ORCIP C-RAN

Development of a Remote Radio Unit and Optical

Access Link for the ORCIP C-RAN Testbed

Dissertação apresentada à Universidade de Aveiro para cumprimento dos requisitos necessários à obtenção do grau de Mestre em Engenharia Eletrónica e Telecomunicações, realizada sob a orientação científica do Professor Paulo Miguel Nepomuceno Pereira Monteiro, Professor Associado do Departamento de Eletrónica Telecomunicações e Informática da Universidade de Aveiro e do Doutor Abel Lorences Riesgo, Investigador Auxiliar no Instituto de Telecomunicações.

This work has been hosted by Instituto de Telecomunicações Aveiro and partially supported by the European Regional Development Fund (FEDER), through the Regional Operational Programme of Centre (CENTRO 2020) of the Portugal 2020 framework, provided by projects ORCIP (CENTRO-01-0145-FEDER-022141), 5GO (POCI-01-0247-FEDER-024539), SOCA (CENTRO-01-0145-FEDER-000010), RETIOT

(4)

(5)

o júri / the jury

presidente / president Professor Armando Humberto Moreira Nolasco Pinto

Professor Associado com Agregação da Universidade de Aveiro

vogais / examiners committee Professor Fernando José da Silva Velez

Professor Auxiliar na Faculdade de Engenharia da Universidade da Beira Interior (arguente principal)

(6)

(7)

agradecimentos acknowledgments

Em primeiro lugar quero agradecer ao meus orientadores, o Professor Paulo Monteiro e os Doutores Abel Riesgo e Fernando Guiomar pela motivação, ensinamentos e apoio para que nada me faltasse na realização deste trabalho.

Agradeço, também, ao Professor Pedro Cabral e ao aluno de doutoramento João Lucas Gomes, pelos preciosos ensinamentos em Eletrónica de Radio-Frequência e ajuda no laboratório.

Ao João Prata, Paulo Gonçalves e Paulo Santos pela simpatia, profissionalismo e disponibilidade em me auxiliar na assemblagem dos circuitos.

Aos meus amigos e colegas de luta diária nestes últimos cinco anos, Rui, Jorge, Pedro e José. Agradeço-vos a amizade, o tempo partilhado e as distrações que tanto facilitaram o trabalho.

Aos meus pais e irmãos, a quem devo todo o investimento de amor e suor na minha educação, que me permitiu chegar até aqui.

(8)

(9)

Palavras-chave Radio-sobre-Fibra, 4G, Frontend de Radio Frequência, ORCIP.

Resumo As novas gerações de comunicações móveis trazem novos desafios para o seu transporte. O suporte a múltiplos serviços, a grande largura de banda e baixa latência, são os requisitos mais importantes.

Esta dissertação aborda a possibilidade do uso de transmissão analógica de radio-sobre-fibra (RoF) na rede de acesso da infraestrutura ORCIP. São estudados e caracterizados transmissores óticos SFP de baixo custo, modificados para transmissão de sinais analógicos. Adicionalmente, é projetada uma unidade de rádio remota contendo um frontend de radio frequência e os transmissores óticos analógicos. O frontend de radio frequência é projetado para operar na banda 7 do LTE. A performance

(10)

(11)

Keywords Radio-over-Fibre, 4G, Radio-Frequency Frontend, ORCIP.

Abstract The new generations of mobile communications bring new challenges for their transport. The support for multiple services, the high bandwidth and the low latency are the most important requirements.

This dissertation explores the use of analog radio-over-fibre transmission in the ORCIP testbed. Low cost SFP optical transceivers modified for analogue signal transmission, are studied and characterised. Additionally, a remote radio unit that contains an RF frontend and the optical transceivers is designed. The RF frontend is designed for operation over band 7 of

(12)

(13)

List of figures

1.1 Schematic showing the concept of a) D-RAN and b) C-RAN. . . 2

1.2 Architecture and implementation strategy of the ORCIP infrastructure [14]. 5 2.1 Architecture of a C-RAN using a) A-RoF and using b) D-RoF. . . 8

2.2 Laser transfer function a) and bias circuit b). . . 9

2.3 MZI transfer function a) and physical structure b). . . 10

2.4 Attenuation characteristic of a silica fibre [15]. . . 12

2.5 Optical power fading induced by fibre chromatic dispersion when using modulation RF signals at 10, 15 and 25 GHz. . . 13

2.6 A RoF link as a subsystem of a large radio link. . . 15

2.7 Plot of output power against input power for an analogue RF system, showing the compression point, intercept points, noise level and dynamic range. . . 17

3.1 Schematic showing the functional blocks of a super-heterodyne RF frontend. 19 3.2 Two port network representation of an amplifier. . . 21

3.3 OFDM modulation scheme. . . 24

3.4 Illustration of the PAPR problem. The system has to reach the required peak power but operates in back-off long periods of time. . . 25

3.5 EVM and related quantities. . . 26

4.1 Implementation diagram illustrating the key building blocks of the indoor-outdoor interface enabled by a RoF path extension between the antenna and the indoor laboratory. . . 28

4.2 Original digital SFP transceiver in a) and modified SFP transceiver in b) [27] 29 4.3 Transceiver circuitry modification, in a) there are the modifications performed in the laser, and in b) the modifications performed in the photodetector [27] . . . 29

4.4 SFP evaluation board. . . 30

4.5 Measured EVM with A-RoF link versus RF frequency and laser RF input power [27]. . . 31

4.6 Experimental setup used to measure the lasers frequency response. . . 31

(16)

List of figures

4.8 Experimental setup used to measure the photodetectors frequency response. 33

4.9 Photodetectors conversion gain a) and output return loss b). . . 34

4.10 Experimental setup used to measure the fibre frequency response. . . 35

4.11 Power gain of a 5 Km fibre span. . . 35

4.12 Detailed architecture and main requirements of the proposed RRU RF frontend. . . 36

4.13 ADL5523 functional diagram a) and evaluation board b). . . 37

4.16 Signal coupler evaluation board. . . 38

4.17 Duplexer evaluation board. . . 39

4.18 Antenna picture a) and antenna without the cover showing the radiating circuit b). . . 39

4.19 LNA evaluation board equivalent circuit. . . 40

4.20 LNA optimal input impedance (inside the blue circle) for the best NF. . . . 40

4.21 LNA forward gain a) and input and output return loss b). . . 41

4.22 LNA NF measured. . . 42

4.23 VGA evaluation board equivalent circuit. . . 42

4.24 VGA forward gain a) and input and output return loss b). . . 44

4.25 VGA NF measurement. . . 45

4.26 Envelope detector board equivalent circuit. . . 45

4.27 Input return loss of the envelope detector evaluation board. . . 46

4.28 VRMS measured as function of RF input power and frequency. . . 46

4.29 Signal coupler board S-parameters. In a) are the transmission coefficients between the input, output and coupled ports. In b) are the reflection coefficients at each port. . . 47

4.30 Duplexer board S-parameters. In a) are the transmission coefficients between the antenna, transmission and reception ports. In b) are the reflection coefficients at each port. . . 48

4.31 Test-set of the RF frontend assembled with the evaluation boards. . . 49

4.32 Measured gain of the uplink path. . . 50

4.33 Stability factor of the uplink path. . . 50

4.34 NF of the uplink path. . . 51

4.35 Measured gain of the downlink path. . . 51

4.36 Stability factor of the downlink path. . . 52

4.37 NF of the downlink path. . . 52

4.38 Fipy micro-controller with extension board. . . 53

4.39 Diagram showing the control signals flow of the RF frontend. . . 54

4.40 Diagram showing the remote operation and monitoring scheme of the RF frontend using MQTT protocol. . . 55

4.41 Graphical interface developed for the remote terminal. . . 56

(17)

List of figures

5.1 Experimental setup used to characterise the RF frontend with a 4G-LTE signal. . . 58 5.2 Experimental setup used to characterise the complete A-RoF path extension

with a 4G-LTE signal. . . 58 5.3 LTE signal EVM (a) and channel power (b) after uplink amplification for

64, 16 and 4 QAM modulation formats. Also in (a), is shown the EVM standard limits for the modulation schemes used. . . 59 5.4 LTE signal EVM (a) and channel power (b) after downlink amplification

for 64 QAM modulation format. Also in (a), is shown the EVM standard limit for the modulation scheme used. . . 61 5.5 LTE signal EVM (a) and channel power (b) after the uplink A-RoF

transmission for 64, 16 and 4 QAM modulation formats. Also in (a), is shown the EVM standard limits for the modulation schemes used. . . 62 5.6 LTE signal EVM (a) and channel power (b) after the downlink A-RoF

transmission for 64 QAM modulation format. Also in (a), is shown the EVM standard limit for the modulation scheme used. . . 63

(18)

(19)

List of tables

3.1 Communication standard’s modulation, PAPR, data rate and bandwidth. . 25

3.2 EVM limits for each modulation format. . . 26

4.1 Optical SFP transceivers main specifications. . . 30

4.2 Components used in the LNA matching networks. . . 40

(20)

(21)

List of abbreviations

3G Third Generation

3GPP 3rd Generation Partnership Project 4G Forth Generation

5G Fifth Generation AC Alternating Current ADC Analog-Digital Converter ADS Advanced Design System APD Avalanche Photodiode A-RoF Analog Radio-over-Fibre AWG Arbitrary Waveform Generator BBU Base Band Unit

CO Central Office

CPRI Common Public Radio Interface

C-RAN Cloud and Centralised Radio Access Network DAC Digital-Analog Converter

DC Direct Current

D-RoF Digital Radio-over-Fibre

D-RAN Distributed Radio Access Network E/O Electrical to Optical

eMBB Enhanced Mobile Broadband EVM Error Vector Magnitude GPIO General Purpose Input Output GUI Graphical User Interface IF Intermediate-Frequency IoT Internet of Things IP Internet Protocol LED Light Emission Diode

(22)

List of abbreviations

LNA Low-Noise Amplifier LTE Long Term Evolution

LTE-A Long Term Evolution-Advanced MIMO Multiple-Input Multiple-Output MMF Multi-Mode Fibre

mMTC Massive Machine Type Communication MPLS Multi-Protocol Label Switching

MQTT Message Queuing Telemetry Transport MZI Mach-Zehnder Interferometer

NF Noise Figure

O/E Optical to Electrical

OFDM Orthogonal Frequency Division Multiplexing OOK On-Off Keying

ORCIP Optical Radio Convergence Infrastructure for Communications and Power Delivering

PA Power Amplifier

PAPR Peak-to-Average Power Ratio PCB Printed Circuit Board

PIN Positive-Intrinsic-Negative

QAM Quadrature Amplitude Modulation QPSK Quadrature Phase-Shift Keying RAN Radio Access Network

RAT Radio Access Technologies RF Radio Frequency

RMS Root Mean Square RoF Radio-over-Fiber RRU Remote Radio Unit SDN Software Defined Network SDR Software-Defined Radio SFP Small Form-Factor Package SMF Single-Mode Fibre

SNR Signal to Noise Ratio UE User Equipment

URLLC Ultra-Reliable Low-Latency Communication VGA Variable-Gain Amplifier

VNA Vector Network Analyser

(23)

List of abbreviations

VOA Variable Optical Attenuator VSA Vector Signal Analyser

(24)

(25)

CHAPTER

1

Introduction

T

user has induced a huge stress in the communication provider infrastructures.he explosion of mobile applications and the extensive adoption by the end Accordingly to the latest visual network reports from Cisco Systems [1, 2], the number of devices connected to the network in 2022 could reach 12 billion. This number includes smart and non-smart devices spread across all the wireless connection technologies. Also, as the mobile devices get smarter, the more data they produce. As a consequence of the increased number of device connections, the amount of traffic exchanged in the network is expected to reach 70 exabytes per month in 2022 compared to the 30 exabytes per month in 2019. In order to enable the full potential of these smart devices, new broadband network technologies appeared, namely the Third Generation (3G) in the early 2000’s and later on the Forth Generation (4G) or Long Term Evolution (LTE) in 2008 [3]. The first one, initially intended to provide support for location-based services, mobile television and video on-demand, achieved its full development in the release 7 from 3rd Generation Partnership Project (3GPP) consortium [4]. 3G mobile communications enable to theoretically downstream and upstream transmission rates of 42.2 Mbit/s and 22 Mbit/s respectively, providing compatibility with Multiple-Input Multiple-Output (MIMO) antennas and high modulation orders, up to 64 Quadrature Amplitude Modulation (QAM). However, it fell short of the requirements for LTE, which demanded Gigabit stationary reception speeds that could be improved with MIMO, in order to provide high-definition live streaming capabilities [5]. Benefiting from the adoption of the Orthogonal Frequency Division Multiplexing (OFDM) scheme to exploit frequency selective property of the channel, it can deliver variable bit rate by assigning different sub-channels to different users [3]. Nowadays, the advent of Fifth Generation (5G), brings new challenges in the development of Radio Access Networks (RANs) physical infrastructure. Among the numerous innovations of 5G, it can be noticed the use of new wave-forms, the support to Enhanced Mobile Broadband (eMBB) with 10× higher capacity and the definition of Ultra-Reliable Low-Latency Communication (URLLC), imposing very stringent requirements in terms of latency and reliability [6]. Also, it is intended

(26)

Chapter 1. Introduction

that the 5G responds to the rising of Internet of Things (IoT), allowing Massive Machine Type Communication (mMTC), in which millions of devices are connected to the network through low-bandwidth channels. In order to embrace new demands, the need to develop a new and future-proof RAN architecture arises [7–9]. In the following, the issues regarding the actual RANs are discussed, and the new RAN typologies to support the next generation Radio Access Technologies (RAT) are presented.

1.1 Motivation

Current mobile broadband services like 3G and 4G are based in a Distributed Radio Access Network (D-RAN), similar to the one presented in Figure 1.1a).

BBU

Backhaul

RF Equipment:

- DACs and ADCs - RF Ampliﬁers - Filters - Mixers

Antennas

RRU = Remote Radio Unit BBU = Base Band Unit CO = Central Oﬀice RRU Core Network a) BBU BBU BBU RRU RRU Core Network CO Backhaul Fronthaul RRU b)

Figure 1.1: Schematic showing the concept of a) D-RAN and b) C-RAN.

The main blocks of this architecture are: the Remote Radio Unit (RRU), the Base Band Unit (BBU) and the fronthaul and backhaul links. The RRU is responsible for transceiving Radio Frequency (RF) signals from and to the User Equipment (UE), amplification of signal power and analogue/digital-to-digital/analogue conversion. The BBU is composed by the digital base band processing circuitry and connects to the core network through a backhaul link, usually using Internet Protocol (IP)/Multi-Protocol

(27)

Label Switching (MPLS) over a optical fibre connection. Signal transmission between the network elements such as the BBU and RRU, relies on the fronthaul link. It is made by a relatively short digital serial link using mainly the Common Public Radio Interface (CPRI) protocol over Digital Radio-over-Fibre (D-RoF) or coaxial cables, as an effort to ensure waveform transparency as well as cost-effectiveness [9]. Although, with the need for a single RRU to support multiple RATs, the use of CPRI already requires an immense bandwidth aggregation on the fronthaul link. For instance, in the most recent flavor of LTE, the Long Term Evolution-Advanced (LTE-A), an channel bandwidth could reach 100 MHz and requires a CPRI-equivalent bit rate of approximately 5 Gbit/s. Therefore, a link that can transport four channels requires a total bit rate of 20 Gbit/s, even without considering additional overhead. Currently, optical transmission systems achieve bit rates above 10 Gbit/s using multiple wavelengths, increasing the cost of the link as result of the mux and demux functions [10]. In the case of an electrical connection the consequences are more severe, as the coaxial cables have high power losses that increase along with frequency, resulting in a tremendous energetic inefficiency. Meanwhile, in the upcoming 5G, networks are envisioned to be heterogeneous where millimetre-wave small cells coexist within larger cells, resulting in a massive deployment of RRUs [9]. To prevent the proliferation of BBUs and ensure a reasonable scalability regarding the infrastructures cost and spatial efficiency, attention is being given to the concept of C-RAN, depicted in figure 1.1 b). This concept aims towards smaller cell sites where the BBUs are remotely co-located in a secure Central Office (CO) in contrast with the traditional arrangement where each BBU is located at the base of the cell tower. Hence, the processing resources can be centralised, in a BBU pool, and dynamic allocated depending on the load of the network. However, C-RAN with respective denser RRU network makes CPRI based fronthaul even more prone to flexibility and bandwidth limitations. New solutions are being investigated in order to address the aforementioned problems, such as: the use of milimeter-wave links [11]; the use of multicore fibres along with Wavelenght Division Multiplexing (WDM) [12]; the use of Ethernet protocol over fibre [13]. Among them, Analog Radio-over-Fibre (A-RoF) signal transmission is generating a lot of interest due to its inherent advantage of making the fronthaul link simpler and completely transparent. The A-RoF implementation helps in shifting the expensive high resolution Analog-Digital Converters (ADCs) and Digital-Analog Converters (DACs) to the BBU pool. Subsequently, the antennas with RF amplifiers, Optical to Electrical (O/E) and Electrical to Optical (E/O) converters are the only components left in the RRU [9]. However, a reliable communication is highly dependent on the opto-electronic components performance to overcome the presence of nonlinearities and inherent lower noise immunity. In the next chapter, the main implementations of Radio-over-Fiber (RoF) will be discussed and compared in more detail. In addition, the impact of the opto-electronic component characteristics in the link performance will be thoroughly evaluated.

(28)

1.2 Objectives

The development of new 5G solutions will inevitably coexist with the current 4G-LTE systems, which are still the key enabler for high-speed mobile communications. Therefore, there is a need to design a flexible testbed that can be scalable, cost-effective and backwards compatible, based on a C-RAN architecture. Also, it should provide support for the new radio technologies such as massive MIMO, beam-forming, multi-band and carrier aggregation. Such testbed will provide an initial platform to test, optimise and validate the aforementioned solutions, potentially reducing the effort required for commercial 5G services provisioning. Several works are being developed in this scope, namely 5GinFire1,

COSMOS 2 and ADRENALINE 3 testbeds. Although, there are some common issues

regarding these testbeds that need to be mentioned: 1) the fronthaul link connection between the BBU pool and the RRUs is based on D-RoF using CPRI protocol. This implementation relies heavily in WDM to provide enough bandwidth; 2) the RRUs are completely of-the-shelf products, which have associated high capital costs as high software license fees. 3) a consequence of the previous aspects, the customisation of the platform, with respect to the fronthaul link and the RRU, is limited. As an overall consequence of these issues, these testbeds become more adequate for Software Defined Network (SDN) and application development.

In order to tackle these problems, it is being developed in the Instituto de Telecomunicões of Aveiro the Optical Radio Convergence Infrastructure for Communications and Power Delivering (ORCIP) testbed [14]. ORCIP aims to provide a platform for the development of new mobile fronthaul networks based on a C-RAN architecture. The generic architecture of the ORCIP testbed is depicted in Figure 1.2. This testbed comprises a set of RRUs deployed over the University of Aveiro Campus and connected to a central laboratory by an A-RoF link. The central laboratory is located inside Instituto de Telecomunicações and it contains all the baseband processing power related to the BBU and the connection to the core network. In addition, the central lab provides the central processing management of the outdoor testbed infrastructure, including RF frontends and optical fibre transceivers. This hybrid in-lab and on-campus approach guarantees a more realistic assessment of the performance of the network in real scenarios. Also, the use of in-house made hardware, along with off-the-shelf commercial equipment, enables the possibility to test new solutions on the outside (RRU side), as well as in the lab (BBU) side without any dependence between them.

In this dissertation work, it is proposed the design and implementation of the A-RoF link connecting the central lab to the RRU. Also, it is intended to design and implement a simple RRU with an RF frontend, antenna and optical transceivers, for operation in 4G over band 7. It is important to stress out that, a Power Amplifier (PA) will not be used in

1 https://5ginfire.eu/university-of-bristol-5g-testbed. 2 https://www.cosmos-lab.org. 3_{http://networks.cttc.es/ons/adrenaline.} 4

(29)

Figure 1.2: Architecture and implementation strategy of the ORCIP infrastructure [14]. the RF frontend, since it is a huge source of non-linearities as well as power consumption and these issues need to be treated with special care. Also, the purpose of this RF frontend is to be employed in a dense network of RRUs with a limited coverage range, usually from 10 to 20 meters. Therefore, the necessary signal power to be transmitted can be achieved without a PA.

1.3 Contributions

The main contributions of this dissertation can be summarised as follows:

1. Analysis of the current mobile network architectures limitations and the need for evolve towards a C-RAN concept.

2. Study and comparison of the A-RoF and D-RoF approaches for future high-capacity communication networks.

3. Characterisation of modified digital Small Form-Factor Package (SFP) optical transceivers for analogue signal transmission.

4. Design of a 4G-LTE RF frontend for operation over band 7, using low-cost commercial components. Resulted in a paper published, as first author, in the conference ConfTele2019 proceedings, with the title "On the Design and Optimisation of an RF Frontend for a Multi-RAT Optical Access Testbed".

5. Implementation of a C-RAN network for the ORCIP tesbed. Resulted in a workshop presented as co-author, in the European Conf. Optical Communications (ECOC 2020) held in Dublin, Ireland, with the title "Optical Network Testbed for Beyond 5G".

(30)

1.4 Dissertation Outline

In order to accomplish the established objectives, this dissertation is organised into six chapters.

To contextualise the problem of using D-RoF in current RAN fronthauls, Chapter 2 provides a condensed review of all RoF transmission techniques and a fair comparison between them, regarding infrastructure complexity, implementation cost and overall performance.

In Chapter 3, a typical architecture of an RRU RF frontend is studied. It is explained the propose of each component of the architecture and the metrics used to evaluate its performance and optimise its design. Also, as the RF frontend project heavily depends on the signals to be transmitted/received, a brief review of the current signals used in 4G systems is presented.

The Chapter 4, addresses the structure of the A-RoF link developed for the ORCIP testbed. The use of modified digital SFP transceivers as analog ones is described and their performance is assessed accordingly to the figures of merit stated in Chapter 2. The influence of fibre lenght on the link response is also tested. The last part of this chapter is dedicated to the RF frontend design and implementation. This chapter includes the review of all the components used and presents the design steps used to optimise the RF frontend.

With the aim to implement a functional fronthaul link and RRU prototype, Chapter 5 presents the validation tests made in-lab, using a real 4G LTE signal.

Finally, in Chapter 6, the main conclusions and suggestions for futures work are summarised.

(31)

CHAPTER

2

Radio-over-Fibre Basics

T

transmission concept. In the following are presented the most used RoF transporthe purpose of this chapter is to present an overview of the RoF signal techniques, namely D-RoF and A-RoF. Both approaches are compared regarding the components used, the deployment cost and the performance achieved. The figures of merit that characterise RoF links are explained too. Ultimately, this aims to show that A-RoF is an effective solution for feeding broadband access points.

2.1 The Concept of Radio-over-Fibre

RoF refers to the principle of transmitting a radio wave over a fibre link, using the radio wave to modulate a optical carrier. On the contrary of traditional optical fibre communication links, in which signal modulation is typically performed in baseband, it is worth mentioning that RoF is associated with the direct optical modulation of a passband RF signal [10, 15]. Therefore, it should be defined as A-RoF links, those where the optical modulation is sufficiently small, and therefore a small signal analysis is possible. In contrast, D-RoF links use optical modulation where the optical source is 100% turned on or off, generally called On-Off Keying (OOK) modulation.

2.1.1 Analogue versus Digital Radio-over-Fibre

Figure 2.1 shows diagrams of an A-RoF and a D-RoF link, which can be used to understand the main advantages and disadvantages of each approach. In Figure 2.1 a), the optic link is realised using A-RoF. For the downlink (signal transportation between the CO and RRU), the baseband data modulates the carrier signal that has the frequency,

fc , producing the final RF signal determined by the adopted wireless technology. Any

images resulting from the mixing process are removed by filtering. Then, through an E/O conversion process, the RF signal modulates an optical carrier to be transmitted in a fibre link. At the RRU, the RF signal is recovered by an O/E converter, amplified and sent to the antenna. The reverse process occurs for the uplink path. Figure 2.1 b) shows

(32)

Chapter 2. Radio-over-Fibre Basics DAC ADC BPF BPF IF IF BPF BPF RF RF E-O O-E Fibre-Optic Ne two rk BPF BPF O-E E-O AMP AMP DSP DSP a) DSP DSP DAC ADC E-O O-E Fibre-Optic Ne two rk O-E E-O AMP AMP DAC ADC IF RF IF RF BPF BPF b)

Figure 2.1: Architecture of a C-RAN using a) A-RoF and using b) D-RoF.

the currently used alternative to the A-RoF technique. In the D-RoF configuration, the baseband signal modulates directly the optical carrier. At the receiver, the O/E converter recovers the baseband signal, then a DAC converts the digital signal to the analog domain to be upconverted to the final RF carrier. After filtering, the RF signal is amplified and ready to be transmitted.

With the previous explanation, the main advantages of using A-RoF become clear. One of the most important is the complexity reduction of the RRU since only the O/E or E/O conversion, RF amplification and filtering are required. This enables a centralised control and management of the signals as well as equipment. A single CO could feed several RRUs, sharing the same signal processing circuitry, resulting in easier accessibility and maintenance. Furthermore, the simpler RRUs and shared resources in the CO makes the A-RoF approach more energy efficient. In the D-RoF configuration, each RRU needs expensive high-speed DACs, ADCs and frequency conversion circuitry. As consequence, control and synchronization signals must be used to sync the converters resulting in a more complex and expensive implementation.

Another advantage of using A-RoF is the spectral efficiency, achieved by using modulation techniques present in wireless systems. For instance, WiFi networks based on the standard IEEE 802.11n transmit at over 100 Mbit/s using a channel bandwidth of less than 40 MHz. On the other hand, a 100 Mbit/s Ethernet link over fibre uses a channel bandwidth of over 125 MHz [16]. As communications systems start to move for bit rates up to 100 Gbit/s and beyond, there is a reason to use more spectrally efficient modulation schemes.

The third advantage for A-RoF systems is their transparency to modulation and signal formats. This argument can be understood in the following example: a wireless link is being updated to operate at 100 Mbit/s instead of 10 Mbit/s. If both implementations

(33)

Chapter 2. Radio-over-Fibre Basics

of Figure 2.1 are compared, it can be seen that in the A-RoF approach, only the wireless signal generation equipment in the CO need to be upgraded. In the D-RoF approach, the digital processing blocks in the CO and the ADCs and DACs at the RRU would need to be upgraded. This becomes more problematic in a multiple (distributed) transmitter system.

In spite of the broad advantages of A-RoF signal transmission, this technique is essentially analogue, and therefore it is highly dependent on Signal to Noise Ratio (SNR) degradation in the link. In the D-RoF implementation, the link includes error control and signal regeneration, preventing the flow of errors to the wireless signal. Also, digital solutions are manufactured in high volume resulting in lower cost components. The reasons aforementioned have meant that A-RoF links have remained in niche applications so far.

2.2 Modulation Types

Radio-over-Fibre links can be also categorised by the type of RF to optical modulation they use. This modulation can be either direct or external. Note that intensity or amplitude and phase modulation can be made with both techniques. The associated detection processes are direct detection and coherent detection, respectively. Due to its simplicity, intensity modulation (along with direct detection) is the most widely used.

2.2.1 Direct Intensity Modulation

In direct intensity modulation, the driving current of the optical source (usually a laser diode) is varied according to the modulating RF signal. This process is shown in Figure 2.2 b). Bias Current Optical Power Linear Region Threshold Spontaneous Emission (nonlinear) Saturation (nonlinear) Modulating RF Signal Light Output RF Source Bias Current Bias Network Laser a) b)

Figure 2.2: Laser transfer function a) and bias circuit b).

Figure 2.2 a) shows the transfer function of a laser diode. In order to operate at the best performance, the laser is biased at the center of the linear region (stimulated emission), where the RF signal modulates this bias current within a confined dynamic

(34)

range. Excess of current will put the laser in the saturation zone leading to nonlinearities, gain compression and harmonic distortion. Below the threshold current, the laser will be in the cut off region and only spontaneous emission occurs. If the input RF signal gets lower than the threshold value of the laser the output optical signal will be clipped.

The simplicity of direct modulation comes at a cost. First, the laser diodes have limited modulation frequency. Commercial available lasers could handle up to 20 GHz RF signals [17]. Also, direct modulation offers high RF loss due to the poor O/E conversion. The modulation gain is given by the slope of the P-I curve (refer to Figure 2.2 a)). In addition, impedance mismatch between the laser and the transmission line that carries the input RF signal increases the loss. Foward-biased lasers have low input impedances, typically in the range of 2-3 Ω, whereas, the transmission line have 50 Ω line impedance. Therefore, the impedance matching circuitry has a big impact in the losses. In that sense, reactive matching networks are preferred because of their low loss properties. However, they have reduced bandwidth. At last, laser intensity modulation is also associated with chirp effect, that causes unwanted frequency modulation. The chirp generates amplitude distortion and exacerbates the fibre chromatic dispersion [18].

2.2.2 External Intensity Modulation

In this case, the laser is driven by a constant bias current, generating a continuous wave light. Then the optical output is intensity modulated by an external optical modulator. This eliminates the issues related to laser chirp, saturation and clipping. The most widely used external modulator is the Mach-Zehnder Interferometer (MZI).

The MZI has very high bandwidth (over 50 GHz) and is capable of handling high continuous optical power. On the other hand, the MZI is expensive and requires a robust mechanical configuration. Moreover, the MZI is a nonlinear device and it could generate harmonics depending on the bias level. This nonlinear nature limits the linear range of operation. Therefore, it requires a biasing level to be done at half power point, as shown in Figure 2.3 a). Optical Power Modulating RF Signal Light Output CW Laser RF Source Bias Network 1 x 2 Splitter Combiner2 x 1 Electrodes a) b)

Figure 2.3: MZI transfer function a) and physical structure b).

The operation on the MZI is depicted in Figure 2.3 b). The continuous light wave from

(35)

a laser is divided equally between the two arms. In one of the arms, an external electric field is applied on the electrodes imposing a small change of the refractive index of the propagation medium. Thus, the optical field suffers a phase shift, ∆φ , that is equivalent to a delay. In the other arm, the optical field remains with the same phase. When combined at the 3 dB power combiner, the phase modulation is converted to intensity modulation. The output will have an intensity peak when ∆φ = 0 and lowest (or zero) intensity when ∆φ = π. Therefore, the MZI can be used for both digital and analog modulation.

2.3 Fibre Channel

Nowadays optical fibre is the most attractive option for RF signal transmission. It has low attenuation and huge bandwidth. Optical fibres can be classified accordingly to the mode of propagation. Single-Mode Fibre (SMF) is designed to allow only the fundamental mode to propagate. On the opposite, Multi-Mode Fibre (MMF) allows the propagation of multiple modes of the same wavelenght. As a consequence, it has large modal dispersion caused by the different group velocities associated with each mode. This causes the fading and spreading of the optical signal along the fibre, therefore limiting the length of the link and the transmission capacity. Therefore, MMFs are usually employed in short-range applications, using low cost Light Emission Diodes (LEDs) as optical power source. For long-reach high capacity links SMFs have been the way to go, due to theirs low dispersion. Therefore, the following study will be focused on them. It will assess the most important aspects of the fibre when designing a RoF link.

2.3.1 Attenuation

The first characteristic that makes the optic fibre so attractive is its low attenuation. This is very important since it increases the power budget and it will have an influence on the wireless link coverage.

The attenuation characteristic of the silica fibre is shown in Figure 2.4 as a function of the optical wavelenght. Rayleigh scattering, absorption due to impurities and the intrinsic absorption of silica are the main power loss causes. It can be seen that Rayleigh scattering dominates at shorter wavelenghts. The two peaks between 1.2 and 1.4 µm are caused by impurities in the fibre. Nevertheless, there are already on the market fibre manufacturers with solutions where those loss peaks are not present [19]. Above 1.6 µm the intrinsic property of silica will absorb the optical power. The lowest attenuation of 0.2 dB/km is achieved within the 1550 nm wavelength region.. To further emphasise the importance of optical losses, it should be referred that any optical power attenuation is doubled (in dB scale) in the electrical domain. The reason is that electric power (at the laser or photodetector) is proportional to the square of the driving/detected current whereas the optical power is directly proportional to the driving/detected current [15]. This topic will be again discussed when the photodetectors characteristics are studied.

(36)

Figure 2.4: Attenuation characteristic of a silica fibre [15].

2.3.2 Chromatic Dispersion

Although, fibre provides signal transmission with low losses, dispersion effects can be more significant than attenuation. In SMFs, the chromatic dispersion is almost zero in the 1310-nm band, however the 1550-nm band offers the lowest attenuation. So, depending on the application a compromise must be made between dispersion and attenuation. In order to understand the effect of chromatic dispersion on RF signal transmission, the transfer function of the fibre will be studied. The following expression represents the transfer function of the fibre, in the frequency domain, considering narrow linewidth lasers [20]:

HSM F (f) = e

jπDδ2_{o (f −fo)}

L c, (2.1)

where D is the chromatic dispersion parameter in ps/(nm.km), λ0is the optical wavelenght

and f is the frequency of the radio signal. Also, L is the fibre lenght in km and c is the speed of light in vacuum.

It can be seen that the phase of the RF signal changes as it propagates through the fibre. Since the modulated optical signal will consist of two modulation sidebands on each side of the optical carrier, the phase change of the two sidebands may cancel each other out. This will result in fading of the signal power detected by the photodiode. The proof of this sideband cancellation effect can be seen as follows.

Let the modulating RF signal s(t) be a real subcarrier:

s(t) = cos(2πfct). (2.2)

The optical field at the output of the laser (which is amplitude modulated), is

eo(t) = [1 + ms(t)]ej2πfot, (2.3)

(37)

where m is the modulation depth of the laser. Performing the Fourier transform of this waveform:

Eo(f) = δ(f − fo) +

m

2[δ(f − (fo+ fc)) + δ(f − (fo− fc)))]. (2.4)

This expression represents the spectrum of an optical carrier and sidebands δ(f −(f0+ fc)) and δ(f−(f0−fc)). The optical signal at the end of the fibre is obtained by multiplying

chromatic dispersion transfer function of equation (2.1) by the frequency-domain optical signal of equation (2.4). Then applying the square law detection at the photodetector, the received RF power can be written as [20]:

PRF(L) ∝ cos2 " πDLλ2_of2 c # . (2.5)

Equation (2.5) shows the received RF power variation as a function of chromatic dispersion parameter, RF signal frequency and fibre length. A simple simulation of this power fading for a single RF carrier is shown on Figure 2.5, using an optical wavelenght of 1550 nm for signal transmission and a dispersion parameter of 17 ps/(nm.km).

0 2 4 6 8 10 12 14 16 18 20 −40 −30 −20 −10 0 Fibre Lenght (Km) Relativ e Receiv ed P o w er (dB) fRF= 10 GHz fRF= 15 GHz fRF= 25 GHz

Figure 2.5: Optical power fading induced by fibre chromatic dispersion when using modulation RF signals at 10, 15 and 25 GHz.

It can be seen that the power fading occurs for shorter fibre lengths as the RF signal frequency increases. Also, if the bandwidth of the RF signal is large enough, this frequency-dependent power fading effect will lead to frequency-selective attenuation of the modulated signal. Techniques like single-side band modulation are commonly used to overcome this issue. C-RAN fronthaul links have at maximum 20 km of distance from the RRU to the CO, which makes the dispersion effect a minor problem up to a few GHz of

(38)

2.4 Optical Receiver

The photodetector is the last component of the RoF link that performs the O/E conversion. As the modulated optical signal propagates through the fibre, it is attenuated, distorted and contaminated with noise. For this reason, it is expected that the receiver detects the RF modulated signal with a given SNR. Hence, the receiver sensitivity can be defined as the mean optical power that must arrive at the photodetector to guarantee a desired SNR for a given bandwidth.

Another important characteristic of a photodetector its responsivity. The responsivity represents the ratio of the optical power (W) received and the resulting output current (A). It can be written as:

<= ηqq

hfo

, (2.6)

where ηq is the quantum efficiency, q is the electron charge, h is Planck’s constant and fo

is the optical signal frequency.

The quantum efficiency can be expressed as the ratio between the number of absorbed photons and the number of emitted electrons. This relation is almost equal to a unit at low frequencies, however high-bandwidth detectors suffer from low quantum efficiency (52% at 100 GHz [21]) and therefore low responsivity. Moreover, the increased bandwidth inherently limits the optical power handling of the photodetector. Under those circumstances, the trade-off between input optical power and linearity must be ensured.

As discussed before for lasers, the matching between the photodetector output impedance and the 50 Ω impedance used in RF systems plays an important role. In this case, the photodiode is reverse biased, providing an high output impedance. Again, the preferred approach is using low loss reactive matching networks.

2.4.1 PIN and Avalanche Photodetectors

The most used types of optic receivers are the Positive-Intrinsic-Negative (PIN) and the Avalanche Photodiode (APD). The PIN diodes are cheap, robust and have acceptable noise performance, which is the reason for their widespread use in optical communications. As the name implies, the APDs have a self-multiplying mechanism that results in internal gain. The drawback of this gain is the resulting noise that comes from the random multiplying process. APDs also require high bias voltages to trigger the multiplication process, which requires additional circuitry.

(39)

2.5 Radio-over-Fibre Performance Metrics

2.5.1 Small-Signal Analysis

Radio-over-Fibre links can be characterised as typical RF systems, as the input and output of the link are RF signals, as shown in Figure 2.6.

Radio-over-Fibre Link

E-O

Converter Optical Link O-EConverter

Figure 2.6: A RoF link as a subsystem of a large radio link.

The link gain can be defined as the ratio between the RF output power and the RF input power. Because of the non-ideal conversion efficiencies of the E/O and O/E converters, the link gain will be affected by them. Thus, the small-signal gain Gd of an

directly modulated RoF link can be defined as [22]:

Gd= Pout Pin = η2_L<2 L2 opt Zout Zin , (2.7)

where Pout is the output RF power, Pin the input RF power, ηLthe laser slope efficiency

(W/A), < is the responsivity (A/W) and Lopt is the loss of the optical link. Zin and Zout

are the input and output impedances respectively of the RoF link. In the case of external modulated links using MZI, the gain Ge is given by [22]:

Ge= PoptηM< LoptLM !2 Zin Zout , (2.8)

where ηM is the slope efficiency of the modulator at the bias point (V−1), Popt is the

optical power at the input of the modulator and LM is the modulator optical insertion

loss. The MZI slope efficiency is given by [22]:

ηM =

πcos φ

2Vπ

, (2.9)

where Vπ is the modulator switching voltage and φ represents the modulator bias point

relative to the quadrature bias (φ = 0).

From equations (2.7) and (2.8) it can be noticed the influence of RF signal frequency in the RoF link gain mainly due to laser/MZI slope efficiencies, photodetector responsivity and impedance mismatch. The effect of dispersion (signal fade/loss) will also have frequency dependence.

Every component in the RoF link adds noise to the system and impacts in the minimum power signal that can be detected by a receiver. This noise is mostly related to the

(40)

temperature and is known by thermal noise. So, every component can be characterised by the Noise Figure (NF) parameter. The NF expresses the ratio of the SNR at the output to the SNR at the input in a logarithmic form [23]:

NF = 10 logSNRout

SNRin

, (2.10)

being SNRout the SNR at the output of the device and SNRin the SNR at the input of

the same device.

Real components have always NF values above 0 dB, which means they always reduce the SNR of the signal that passes through them. For a passive device it can be demonstrated that its NF value is equal to its insertion loss [24]. In case of active devices the NF depends in the impedance of the signal source and it will be studied in Chapter 3.

An RoF link is composed by several stages cascaded together, increasing the noise power and therefore reducing the SNR. In order to calculate the noise contribution of the whole n-stage system NFtot, one should apply the Friis’s formula [23] :

N Ftot= 10 log F1+ F2−1 G1 + Fn−1 G1G2...Gn−1 , (2.11)

where F is the noise factor, which is equivalent to 10NF/10 and G is the numerical power

gain, which is equal to 10A/10, where A is the power gain in dB.

As shown in equation (2.11), the impact in NF is mostly related to the first element of the system cascade. So, for the best system noise performance, the first stage should have a low noise figure and the highest possible gain. In a RoF link (refer to Figure 2.6), this is done by adding low-noise pre-amplifiers prior to the E/O conversion. However, excessive pre-amplification could drive the optical link into the nonlinear region. Reducing the excess of noise is crucial specially in the uplink path (refer to Figure 2.1a)) where the signal power received at the antenna is considerably low.

2.5.2 Large-Signal Analysis

In the previous analysis the RoF link was characterised assuming that it was linear. However, real components may become nonlinear when excited with high power levels. The most detrimental effects caused by nonlinearities are gain compression, generation of undesired frequency components or even complete failure. These effects can lead to signal losses/distortion and interference with other radio channels.

Figure 2.7 represents a typical plot of output power against input power for an analog system.

On a logarithmic scale, the slope of the fundamental component at the output is generally unitary, but will deviate as the devices in the link saturate. This effect is called gain compression or saturation. To quantify the linear operating range of the system, the P1dB is defined as the power level for which the output power decreases 1 dB from

(41)

Chapter 2. Radio-over-Fibre Basics Output P ow er (dBm ) Input Power (dBm) 1 dB Compression Point Intercept Points Noise Level 2nd order product Fundamental component 3rd order product SNR Signal/ Distortion Dynamic Range

Figure 2.7: Plot of output power against input power for an analogue RF system, showing the compression point, intercept points, noise level and dynamic range.

the linear characteristic. In addition, the second and third order products created by the link nonlinearity are also shown in Figure 2.7. The second order products are measured at twice the fundamental frequency, or at f1 + f2 for a two-tone input with tones at f1

and f2. The third order product is measured also with a two tone input at 2f1 − f2 or

2f2− f1. The latter is the most important as it commonly appears within the bandwidth

of the system causing the spectral regrowth phenomena and interference with adjacent channels. The point at which the linear extrapolation of the third order curve intercepts the linear extrapolation of the fundamental curve is called the IP3. It could be referenced

as an input/output power level of the system.

As receivers require a certain SNR in order to achieve a certain performance level, its dynamic range should be defined regarding the previous analysis. The dynamic range can be expressed as the difference between the minimum input power that ensures a defined SNR and the maximum input power the ensures a defined signal-to-distortion ratio. In other words, the noise affects the lower end of the dynamic range, while the higher end is affected by component nonlinearity.

For RoF links the nonlinear distortion is usually related with the laser, in the transmitter side and with the photodetector in the receiver one. In a directly modulated link, the laser linear region is limited by the threshold and saturation current. In addition, a laser is essentially a optical oscillator and its linear characteristic is only observed under static conditions. When using an MZI for external intensity modulation, a trade off must be done. Quadrature biasing, as shown in Figure 2.3 a), maximises the slope efficiency

(42)

and reduces the second order nonlinearity. On the other hand, using a bias point near the minimum reduces the third/odd order nonlinearity [25]. Photodetector nonlinearity becomes a problem at high optical power levels, in the range of a few milliwatts. Also, the responsivity decreases with increased optical power. As with photodetectors, optical fibre nonlinearity only shows off at high optical power levels, in the range of tens of milliwatt for SMFs [10].

(43)

CHAPTER

3

Radio Frequency Frontend Basics

I

n this chapter a general architecture of a RF frontend used in current wireless systemsis presented. The purpose of each block is explained giving the emphasis to the amplifiers. Also, the design guidelines to optimise amplifiers performance is assessed. As the RF frontend project depends on the signals to be transmitted or received, the characteristics of the signals used in the 4G-LTE systems are described.

3.1 Functional Description

The RF frontend is a subsystem of the radio receiver-transmitter system. The operation of the commonly used super-heterodyne RF frontend is depicted Figure 3.1.

DAC ADC LNA

LNA PA Gain Control Block

Gain Control Block

Antenna R T BPF BPF BPF BPF Downconverter Upconverter RF Oscillator Gain Block Duplexer

Digital Signal Processor

I/Q Demodulator

I/Q Modulator

Figure 3.1: Schematic showing the functional blocks of a super-heterodyne RF frontend. Starting from the left, the antenna is the first and the last element of the RF frontend. It has the function of radiating and receiving radio waves, being the transitional structure between a free-space and a guiding device. Its main characteristics are the bandwidth,

(44)

Chapter 3. Radio Frequency Frontend Basics

the gain and the radiation pattern. The bandwidth of the antenna defines the frequency range where it can receive or transmit radio signals. The gain is the measure of how well an antenna can convert the input power into radio waves in a specified direction. Finally, the radiation pattern represents the antenna radiation properties such as power flux and density, in a spatial domain.

Then, to separate or join the transmission and reception channels a device called duplexer is used. It performs frequency multiplexing allowing disjoint frequency channels to share a common port. In this case, the antenna port. High isolation between the transmission and reception ports must be ensured so that the receiver cannot be desensitised. Also, the insertion loss between the antenna/transmitter port and reception/antenna port should be as low as possible to reduce signal power loss.

In the reception chain (the upper part of the figure), the first component is an Low-Noise Amplifier (LNA). It should provide considerable gain and low NF, in order to minimise the SNR degradation, as the received RF signal has already a low power level. Then, a variable gain block is added so that, depending on the distance from the transmitter to the receiver, the power can be controlled to prevent overloading the following components.

A filter is used to remove unwanted spectral components, that were previously amplified, while providing low insertion loss at the band of interest.

Next, a mixer down-converts the RF signal coming from the filter to an Intermediate-Frequency (IF) using the reference of an oscillator. Typically, the most important characteristics in a mixer are the isolation between ports and suppression of spurious mixing products.

The oscillator should provide a stable frequency reference with the capability of tuning in frequency increments that support the system’s channel bandwidths. In addition, the local oscillator must provide low phase noise and enough power to drive the mixer.

Again, signal is filtered to remove any mixing products left behind. Afterwards, the analogue signal is converted by an ADC to the digital domain, where an I/Q demodulator recovers the I and Q data signals.

The transmission chain (on the lower part of the figure) is the reverse of the reception one. Firstly, the I/Q modulator generates the I/Q signal and a DAC converts it to the analogue domain. An LNA increases the signal power to compensate for the signal losses of the subsequent image-reject filter. The mixer upconverts the baseband signal to the final RF carrier and it is followed by a variable gain block and a PA that give the final power boost to the RF signal before it transmission at the antenna.

3.2 Amplifier Theory

As previously said, most of the frequency conversion and modulation/demodulation made in an RF frontend, is now performed in the digital domain. This is due to the decreasing cost of high sampling frequency ADCs/DACs and the increasing performance

(45)

of digital signal processors. Nevertheless, some operations still need to be performed in the analogue domain, such as, signal amplification. In the following, a set of metrics commonly utilized to evaluate the performance of RF amplifiers will be briefly reviewed.

3.2.1 Power Gain

When designing an amplifier, in terms of gain, the power gain is the most important to consider. However, there are several ways to define the gain of an amplifier. In order to clarify the different ways to express gain, an amplifier schematic, characterised by its scattering parameters, with respective load and power sources is shown in Figure 3.2.

Figure 3.2: Two port network representation of an amplifier.

The power gain G is the ratio of power dissipated in the load PLto the power delivered

to the input of the amplifier, Pin.

The available power gain GA is the ratio of power available from the amplifier Pavn

and the power available from the source Pavs.

At last, the transducer power gain GT is the ratio of the power delivered to the load

PL and the available power at the source Pavs.

As can be seen, the power gain, G, and the available power gain, GA, do not

show dependence on the source and load impedances, respectively. Still, active devices characteristics, such as NF and efficiency, usually depend on the ZL and ZS and must be

accounted for the final amplifier design. Moreover, a perfect conjugate matching assumed in the aforementioned definitions is hard to achieve, therefore they are only applicable in an ideal scenario. Therefore, the most accurate definition of gain that will be used for the rest of this work is the transducer gain GT, defined as [24]:

GT = PL Pavs = |s21|2 1 − |ρ2S| 1 − |ρ₂ L| |1 − ρ_Sρin|2|1 − s22ρL|2 , (3.1)

where, ρin is the reflection coefficient measured at the input of the amplifier when the

output is terminated, given by [24]:

ρin = s11+

s12s21ρL

1 − s22ρL

(46)

3.2.2 Stability

The stability of a system is an important aspect to address when designing RF circuits. Otherwise, an unstable circuit could lead to the system malfunction or damage. This has to be accounted early in the design stage. Oscillation occurs if either the input or output port impedance of a circuit has a negative real part; which means that ρin>1 or ρout>1.

Because ρin and ρout depend on the source and load matching networks, and the latters

are frequency dependent, it is possible for an amplifier to be stable at a frequency region but unstable at another. Therefore, one can define two types of stability: unconditional stability and conditional stability. The first assumes the amplifier stability for all passive source and load impedances. Instead, the conditional stability corresponds to the case when an amplifier is stable for a certain range of passive source and load impedances.

A first metric used to determine the unconditional stability is the Rollet Factor, or

K −∆ test. The test parameters are defined as [24]: K= 1 − |s11| 2_|s 22|2+ |∆|2 2|s12s21| , (3.3) where: |∆| = |s11s22− s12s21|. (3.4)

The unconditional stability is achieved if K > 1 and |∆| > 1. If the S parameters do not satisfy this method, the stability circles must be drawn in the Smith Chart to verify the source and load impedances that correspond to the stable region. The radius, r, for those circles are given by Equation (3.5) for the output and Equation (3.6) for the input. The center impedances, c, are given by Equation (3.7) for the output and Equation (3.8) for the input.

rL= s12s21 |s₂₂|2_{− |}∆|2 (3.5) rS = s12s21 |s11|2− |∆|2 (3.6) cL= (s22 −∆s∗₁₁) |s22|2− |∆|2 (3.7) cS = (s11 −∆s∗₂₂) |s₁₁|2_{− |}∆|2 (3.8)

The previous equations only show the boundary between the stable and the unstable region. It remains to be confirmed if the stable region is inside or outside the circles. So, the stability conditions should be solved choosing a certain load in the Smith Chart. For simplicity, it is usually chosen the center of the Smith Chart, since the respective reflection coefficient is zero. In every circuit design, stability must be ensured in the

(47)

in-band frequencies as well as in the out-of-band ones. If possible, the stability must be tested in large signal conditions. Active devices, like amplifiers, change its characteristics with input power, so stability should be verified for several power levels.

3.2.3 Noise Figure

Another important design consideration for a microwave amplifier is its NF. As previously mentioned, in a receiver it is often required to have an LNA as the first stage, due to its dominant effect in the overall noise performance. Generally it is not possible to obtain low noise and maximum gain, so a compromise must be made. This can be done by calculating constant gain circles and constant NF circles to select the trade-off.

The constant noise figure circles are centred at [24]:

cf =

ρopt

N + 1, (3.9)

where, ρopt is the optimum source reflection coefficient that results in the minimum noise

figure and N is given by:

N = F − Fmin

4RN/Zo

|1 + ρopt|2. (3.10)

RN is the equivalent noise resistance, Fminis the minimum noise factor of the transistor

and Z0 is the characteristic impedance used in the system, commonly 50 Ω. RN and Fmin

are given by the manufacturer. The noise factor F can be expressed as:

F = Fmin+ 4RN

Z0

|ρ_S− ρ_opt|2

(1 − |ρS|2) |1 + ρopt|2

. (3.11)

This equation already shows that the noise figure is only dependent on the source impedance.

The radius of the noise circle can be given by:

rF =

q

N(N + 1 + |ρopt|2)

N+ 1 . (3.12)

For an LNA design, only the constant gain circles at the input have interest because the NF only depends on the source impedance. So, it is implied conjugate matching at the output for maximum power gain. Therefore, the input circle center and radius is given by: cS = gSs∗22 1 − (1 − gS) |s11|2 , (3.13) rS = p 1 − gS(1 − |s11|2) 1 − (1 − gS) |s11|2 , (3.14) where gS is:

(48)

Chapter 3. Radio Frequency Frontend Basics gS = 1 − |ρS|2 |1 − s11ρS|2 1 − |s11|2 . (3.15)

Performing the calculations of the constant noise circles and constant gain circles by hand can be cumbersome. Nowadays, some manufacturers provide complete models of the devices that can be used in a design software, reducing the implementation time.

3.3 4G Signal Basic Characteristics

The appearance of 4G-LTE allowed the provision of real-time services with substantial increase in the transmission capacity and spectral efficiency over previous wireless technologies. This only became possible with the adoption of multi-carrier modulation schemes such as OFDM, shown in Figure 3.3.

Tu

T_u

Fc1Fc2Fc3 f = 1

Data S/P OFDM_Modulation

High bit rate signal M-QAM M-QAM M-QAM M-QAM

Figure 3.3: OFDM modulation scheme.

In OFDM, a high-speed serial data is divided into several parallel streams with lower bit-rate. These substreams are used to modulate multiple parallel subcarriers. Every OFDM waveform is chosen in such a way that mutual orthogonality among subcarriers exist. This makes the OFDM robust against fading in certain frequencies by avoiding transmission in a particular band our reducing the modulation order. Also, multi-path fading and inter-symbol interference impact is reduced with OFDM, as the symbol duration increases for the lower bit-rate subcarriers. Another advantage is the frequency diversity , as different symbols arrive at the receiver on independent orthogonal subcarriers.

3.3.1 Peak-to-Average Power Ratio

The drawback of OFDM modulation is the Peak-to-Average Power Ratio (PAPR) (defined in Equation (3.16)) of the produced signals. This effect, illustrated in Figure 3.4, imposes a limitation on the dynamic range of a system, as it is most of the time operating

(49)

in the back-off region. Anyway, it has to be capable of receive high peak input power without producing distortion. In an RoF link, besides the loss of dynamic range, a loss in the link power budget is noticed, as the devices cannot be excited with the maximum input power. Time Amplitude High Frequency Carrier Peak Power Average Power Envelope

Figure 3.4: Illustration of the PAPR problem. The system has to reach the required peak power but operates in back-off long periods of time.

PAPRdB = 10 log10

Ppeak

Pavg

!

(3.16) To further exacerbate this problem, the latest signal standards tend to have higher bandwidths and consequently more subcarriers, thus increasing the PAPR. Table 3.1 shows the evolution of the telecommunication signals (downlink path) regarding bandwidth, PAPR and data rate.

Table 3.1: Communication standard’s modulation, PAPR, data rate and bandwidth. Standard Modulation PAPR Data Rate Bandwidth GSM (2G) GMSK 0 dB 22.8 kbps 200 kHz EDGE (2.5G) 8-PSK 3.2 dB 59.2 kbps 200 kHz UMTS (3G) QPSK 3.5 - 7 dB 2 Mbps 5 MHz

LTE (4G) OFDM 9 - 12 dB 20 Mbps 20 MHz

3.3.2 Error Vector Magnitude

Currently, wireless signal standards use multi-level and multi-phase modulation formats like M-QAM. In order to assess their quality, the EVM metric is commonly employed. Error Vector Magnitude (EVM) considers all the potential phase and amplitude distortions as well as noise and provides a single measurement figure for determining the quality of a system.

(50)

Using phasors in the I/Q plane, EVM illustrates the reference or ideal symbol vector location and size compared to the actual measured vector, as shown in Figure 3.5. The difference between the positions of the ideal or reference phasor and the actual received phaser is the EVM.

I

Magnitude Error Measured signal Reference signal Phase error Error Vector

{

Q

Figure 3.5: EVM and related quantities.

As it is a time-domain measurement the EVM can be defined as follows: EVM(%) = v u u t P n|yn− xn|2 P n|xn|2 (3.17) where xn is the ideal waveform intended for the nth waveform sample, and yn is the

received waveform.

Depending on the depth of the modulation, maximum EVM values are defined for correct demodulation of the signals. Table 3.2 presents the EVM limits accordingly to the modulation format defined by 3GPP [26].

Table 3.2: EVM limits for each modulation format. Modulation Format EVM limit (%)

QPSK 17.5

16 - QAM 12.5

64 - QAM 8

256 - QAM 3.5

Development of a remote radio unit and optical access link for the ORCIP C-RAN testbed

Bruno Tavares

Brandão

Desenvolvimento de uma Unidade de Rádio

Remota e Rede Ótica de Acesso para a

Infraestrutura ORCIP C-RAN

Bruno Tavares

Brandão

Desenvolvimento de uma Unidade de Rádio

Remota e Rede Ótica de Acesso para a

Infraestrutura ORCIP C-RAN

Development of a Remote Radio Unit and Optical

Access Link for the ORCIP C-RAN Testbed

Table of contents

List of figures

List of tables

List of abbreviations

CHAPTER

1

Introduction

T

1.1

Motivation

1.2

Objectives

1.3

Contributions

1.4

Dissertation Outline

CHAPTER

2

Radio-over-Fibre Basics

T

2.1

The Concept of Radio-over-Fibre

2.2

Modulation Types

2.3

Fibre Channel

2.4

Optical Receiver

2.5

Radio-over-Fibre Performance Metrics

CHAPTER

3

Radio Frequency Frontend Basics

I

3.1

Functional Description

3.2

Amplifier Theory

3.3

4G Signal Basic Characteristics

I

{

Q