Universidade de Aveiro Departamento deElectr´onica, Telecomunica¸c˜oes e Inform´atica 2016
Andr´
e Mariano Silva
Ferreira
Caracteriza¸
c˜
ao em Frequˆ
encia de Rel´
es RF de
Bobina ´
Unica
Universidade de Aveiro Departamento deElectr´onica, Telecomunica¸c˜oes e Inform´atica 2016
Andr´
e Mariano Silva
Ferreira
Caracteriza¸
c˜
ao em Frequˆ
encia de Rel´
es RF com
uma Bobina
Frequency Characterization of Single Coil RF Relays
Disserta¸c˜ao apresentada `a Universidade de Aveiro para cumprimento dos requisitos necess´arios `a obten¸c˜ao do grau de Mestre em Engenharia Eletr´onica e Telecomunica¸c˜oes, realizada sob a orienta¸c˜ao cient´ıfica do Pro-fessor Doutor Pedro Miguel da Silva Cabral, ProPro-fessor Auxiliar do Departa-mento de Electr´onica, Telecomunica¸c˜oes e Inform´atica da Universidade de Aveiro e sob a co-orienta¸c˜ao cient´ıfica do Professor Doutor Jos´e Carlos Es-teves Duarte Pedro, Professor Catedr´atico do Departamento de Electr´onica e Telecomunica¸c˜oes da Universidade de Aveiro.
Dissertation presented to the University of Aveiro for the fulfillment of the necessary requirements to obtain the degree of Master In Electronics and Telecommunication Engineering, developed under the scientific guidance of Dr. Pedro Miguel da Silva Cabral and Dr. Jos´e Carlos Esteves Duarte Pedro, Professors at the Department of Electronics, Telecommunications and Informatics of the University of Aveiro.
O J´uri / The Jury
Presidente / President Professor Doutor Nuno Miguel Gon¸calves Borges de Carvalho
Professor Catedr´atico da Universidade de Aveiro
Arguente Principal / Main Examiner
Professor Doutor Pedro Renato Tavares de Pinho
Professor Adjunto do Instituto Superior de Engenharia de Lisboa
Orientador / Advisor Professor Doutor Pedro Miguel da Silva Cabral
Agradecimentos Em primeiro lugar, gostaria de agradecer `a minha fam´ılia por todo o amor e sacrif´ıcio ao longo destes anos.
Agrade¸co aos meus orientadores Prof. Pedro Cabral e Prof. Jos´e Car-los Pedro por todo o apoio, disponibilidade e valiosas sugest˜oes que tanto contribuiram para este trabalho.
Gostaria tamb´em de agradecer `a TE Connectivity de ´Evora, pela opor-tunidade de realizar parte deste trabalho nas suas instala¸c˜oes, e a todos os seus colaboradores que me ajudaram durante a realiza¸c˜ao do mesmo. N˜ao posso deixar de agradecer `a Universidade de Aveiro, ao Departa-mento de Electr´onica, Telecomunica¸c˜oes e Inform´atica, ao Instituto de Telecomunica¸c˜oes e a todos os seus colaboradores pelas excelentes condi¸c˜oes de trabalho e aprendizagem que me proporcionaram. Por fim um agradecimento especial aos meus amigos, pelo apoio in-condicional ao longo de todos estes anos.
Palavras-Chave Calibra¸c˜ao do VNA, Kit de calibra¸c˜ao personalizado, Medi¸c˜oes em Pla-cas de Teste, Rel´es RF, SOLT, TRL, VNA
Resumo Short-Open-Load-Thru (SOLT) e Thru-Reflect-Line (TRL) s˜ao as duas
t´ecnicas de calibra¸c˜ao mais utilizadas na correc¸c˜ao de erros aplicada em Vector Network Analyzers (VNA). No entanto, alguns requisitos tˆem que ser cumpridos, de modo a permitir a aplica¸c˜ao destas t´ecnicas de calibra¸c˜ao em medi¸c˜oes realizadas em placas de teste. Esta disserta¸c˜ao destina-se `a caracteriza¸c˜ao em frequˆencia de dispositivos inseridos em placas de teste, mais especificamente de rel´es electromecˆanicos de bobina ´unica para aplica¸c˜oes em frequˆencias r´adio (RF) e microon-das. As t´ecnicas de calibra¸c˜ao SOLT e TRL s˜ao inicialmente estu-dadas de modo a perceber os seus requisitos e limita¸c˜oes. De seguida, desenvolve-se e apresenta-se o desenho de standards de calibra¸c˜ao per-sonalizados em placas de teste. Por fim, para cada t´ecnica de cal-ibra¸c˜ao, procede-se `a caracteriza¸c˜ao em frequˆencia de rel´es RF de bobina ´unica e comparam-se os resultados obtidos com as curvas de desempenho t´ıpicas apresentadas pelo fabricante.
Keywords Custom CPWG calibration kit, In-fixture measurements, RF relays, SOLT, TRL, VNA, VNA Calibration
Abstract Short-Open-Load-Thru (SOLT) and Thru-Reflect-Line (TRL) are the
two most common calibration techniques for the error correction ap-plied on Vector Network Analyzers (VNA). These techniques however are not so straight forward when it comes to in-fixture measurements. This dissertation is aimed at the frequency characterization of in-fixture devices, specifically single coil electromechanical relays for radio fre-quency (RF) and microwave applications. The SOLT and TRL cali-bration techniques are first studied to understand the limitations and requirements each technique. Then the design of custom in-fixture calibration standards is developed and presented for each technique. Finally, the single coil RF relays are characterized for each calibration technique and the results are compared with the typical performance presented on the datasheet.
Contents
Contents i List of Figures v List of Tables ix Acronyms xi 1 Introduction 1 1.1 Background . . . 11.2 Motivation and Objectives . . . 2
1.3 Organization and Structure . . . 4
2 Scattering Parameters 7 2.1 Introduction . . . 7
2.2 Reflection Coefficient . . . 7
2.3 N-Port Networks . . . 9
2.4 Scattering Transfer Parameters . . . 11
3 Vector Network Analyzer 13 3.1 Introduction . . . 13
3.2 Internal Block Diagram . . . 14
3.3 Measurement Errors . . . 16
3.4.1 Twelve-Term Error Model . . . 20
SOLT Calibration . . . 22
3.4.2 Eight-Term Error Model . . . 25
TRL Calibration . . . 26
3.5 VNA Models for the Calibration Standards . . . 28
4 In-Fixture Custom Calibration Kit 31 4.1 Introduction . . . 31
4.2 RF Relays from TE Connectivity . . . 32
4.2.1 Internal Structure of the HF Series Relays . . . 34
4.3 Calibration Standards Design and Definition . . . 36
4.3.1 Short Standard . . . 40
4.3.2 Open Standard . . . 42
4.3.3 Load Standard . . . 46
4.3.4 Thru Standard . . . 47
4.3.5 Line Standard . . . 48
4.4 Improvements over the V032K Calibration Kit Design . . . 51
4.5 Experimental custom TRL Calibration Kit . . . 53
4.6 Cost Evaluation . . . 60
5 Frequency Characterization of RF Relays 63 5.1 Figures of Merit of RF Relays . . . 65
5.1.1 Isolation . . . 65
5.1.2 Insertion Loss . . . 66
5.1.3 VSWR . . . 66
5.2 SOLR and TRL Calibration Procedures . . . 67
5.3 Characterization Procedure and Results . . . 70
6.2 Future Work . . . 76
Bibliography 77
A HF3 Series Datasheet 81
B HF3 S Series Datasheet 89
C RO4000 Series Datasheet 95
D Design of the Aluminum Boards 101
E Vishay CH Series Datasheet 107
F CGH35015F Datasheet 119
G PCB Layout of Kit CGH35015F and Test-Fixture 129
List of Figures
1.1 Basic representation of an RF relay integrated on a radio system . . . 2
1.2 E8361C Vector Network Analyzer from Agilent Technologies . . . 3
2.1 Transmission line with the source end at x = l and load end at x = 0 . . . 8
2.2 Signal flow graph representation of the S-parameters of a two-port network 9 2.3 Cascaded two-port networks . . . 12
3.1 Standard block diagram of an N-port vector network analyzer . . . 14
3.2 Simplified test set block diagram of the E8361 VNA used in this work . . . 15
3.3 Systematic measurement errors . . . 17
3.4 Twelve-term error model . . . 20
3.5 85052D 3.5 mm SOLT calibration kit from Agilent Technologies . . . 24
3.6 Classes of DUT . . . 24
3.7 Two-port eight-term error model . . . 25
3.8 TRL calibration kit with two delay lines and a short . . . 26
3.9 Model of an open standard . . . 29
3.10 Model of a short standard . . . 29
3.11 Model of a load standard . . . 30
3.12 Model of a thru/line standard . . . 30
4.1 RF relays under test . . . 32
4.3 Exploded view of an HF3 series relay . . . 34
4.4 Design principle the HF3 series relays . . . 35
4.5 Coaxial test-fixture . . . 36
4.6 Custom SOLT calibration kit . . . 37
4.7 Custom TRL calibration kit with N line standards . . . 37
4.8 Measurement reference plane set by the calibration kits . . . 38
4.9 Cross section of a CPWG transmission line . . . 38
4.10 Aluminum board for Kit IT v2 (on the left) and respective test-fixture (on the right) . . . 39
4.11 Design of a CPW short circuit . . . 40
4.12 Design of the short standard from Kit IT v2 . . . 40
4.13 In-fixture short standard from Kit IT v2 . . . 41
4.14 Design of a CPW open circuit . . . 42
4.15 Design of the open standard from Kit IT v2 . . . 42
4.16 In-fixture open standard from Kit IT v2 . . . 43
4.17 Open standard measured with exaggerated port extension delay . . . 44
4.18 Phase response of the open standard . . . 45
4.19 Design of the load standard from Kit IT v2 . . . 46
4.20 In-fixture load standard from Kit IT v2 . . . 46
4.21 Design of the thru standard from Kit IT v2 . . . 47
4.22 In-fixture thru standard from Kit IT v2 . . . 48
4.23 Design of the line standard from Kit IT v2 . . . 49
4.24 In-fixture line standard from Kit IT v2 . . . 49
4.25 Insertion phase of the line standards . . . 50
4.26 Custom SOLT calibration kit from TE Connectivity (a) and IT (b) . . . . 51
4.27 Improvements over the design of the open standard . . . 52
4.28 Improvements over the design of the short standard . . . 53
4.31 Cross section of a microstrip transmission line . . . 55
4.32 Measurement setup for a CGH35015F GaN transistor . . . 55
4.33 Simulation setup for a CGH35015F GaN transistor . . . 56
4.34 CGH35015F S-Parameters @ VDS = 28V and IDS = 0mA . . . 57
4.35 CGH35015F S-Parameters @ VDS = 28V and IDS = 54mA . . . 58
4.36 CGH35015F S-Parameters @ VDS = 28V and IDS = 106mA . . . 59
5.1 Kit IT v2 . . . 64
5.2 Test-fixture for HF3 and HF3 S series relays . . . 64
5.3 Signal transmission on an RF relay . . . 65
5.4 SOLT vs SOLR calibrations using the V032K calibration kit . . . 68
5.5 SOLR and TRL calibration procedures . . . 69
5.6 Measurement setup for HF3 and HF3 S series relays . . . 70 5.7 Comparison between the typical RF performance, displayed on the datasheet,
and the HF3 53 RF parameters measured using different calibration kits . 71 5.8 Comparison between the typical RF performance, displayed on the datasheet,
List of Tables
2.1 Conversion between S and T parameters . . . 12
3.1 Systematic error terms . . . 21
3.2 Raw parameters measured from the calibration standards . . . 21
3.3 S-parameter matrices of ideal calibration standards . . . 22
Acronyms
ADS Advanced Design System
CPWG Co-Planar Waveguide with Lower Ground Plane
DUT Device Under Test
EDA Electronic Design Automation
GaN Gallium Nitride
HF High Frequency HP Hewlett-Packard IF Intermediate Frequency IT Institute of Telecommunications M2M Machine to Machine NC Normally-Closed
NEP Network Equipment Provider
NIST National Institute of Standards and Technology
NO Normally-Open
R&D Research and Development
RF Radio Frequency
SMA SubMiniature Version A
SMT Surface Mount Technology
SOLR Short-Open-Load-Reciprocal
SOLT Short-Open-Load-Thru
SPDT Single Pole Double Throw
TRL Thru-Reflect-Line
TRM Thru-Reflect-Match
VNA Vector Network Analyzer
Chapter 1
Introduction
1.1
Background
There are more mobile devices globally connected to the internet (7.9 billion in 2015) than there are people on Earth. It is expected that by 2020 there will be 1.5 mobile devices per capita which, together with Machine to Machine (M2M) modules, will sum up to a total of 11.6 billion connected mobile devices [1].
Network Equipment Providers (NEP) invest in the Research and Development (R&D) of new reliable and efficient systems, in order to respond to the increasing number of users. It is by continuously innovating their services and products, that NEP are able to keep up with such a fast changing market.
During the R&D of these systems, it is very useful to use representative models of the devices being used. These models allow engineers to predict the behavior of such devices, or even whole systems, even before production stage. With this in mind, most manufacturers characterize their products and provide the resulting data to designers, though sometimes only upon request.
1.2
Motivation and Objectives
Most telecommunication systems require suitable switching elements to transmit or in-terrupt Radio Frequency (RF) and microwave signals. One of the most commonly used devices to perform this task are electromechanical RF relays, hereinafter known simply as RF relays. Despite being based on a relatively old switching technology, they are still preferred for having high levels of isolation and low insertion loss within a broad range of frequencies. RF relays are also able to carry DC signal superimposed to the RF signal and transmit high power RF signals without distortion [2]. This set of characteristics make RF relays a good RF switching solution to be used on measurement and test equipment, wireless base stations, RF power amplifiers and other wireless infrastructure components [3] (fig. 1.1).
Figure 1.1: Basic representation of an RF relay integrated on a radio system
At radio and microwave frequencies, the size of most RF components is comparable to the wavelength of the signals exciting them. As a result, the integrity of such High Frequency (HF) signals can be affected by [4]:
• Phase Shift - The phase of high-frequency signals may shift as they flow through the component.
• Dissipation - As the frequency of a signal increases so does the power it dissipates in the circuit. This, in turn, may lead to an increase of thermal noise.
• Noise - RF and microwave circuits tend to generate noise that may compromise the integrity of the signal.
• Radiation - It leads to unwanted coupling between circuits, but is also desirable in the design of circuits dedicated to the reception or transmission of RF signal.
• Reflections - Cascading elements of different impedance causes high frequency sig-nal to be partially or even totally reflected, reducing power transmission and overall efficiency.
Fortunately these effects can be measured from RF devices, by using dedicated equip-ment known as Vector Network Analyzer (VNA) (fig. 1.2). The VNA is able to measure a wide range of RF devices, from small discrete components to fully assembled systems. However, due to the diversity of RF devices, VNAs may require the use of different acces-sories to perform different measurements. For instance, devices terminated with different connectors require different cables, connectorless components require test-fixtures to be fixed on, and high power amplifiers require isolators and attenuators to avoid damaging the VNA.
VNAs also allow the user to remove unwanted effects from the measurement setup, through a process known simply as VNA calibration. By using this process the user is able to shift the measurement plane from the ports on the front panel, to the terminals or connectors of the device being measured. There are several calibration techniques to choose from, each best suited for different situations. Choosing an adequate calibration procedure is essential to obtain accurate results.
The main goal of this dissertation is to perform the in-fixture characterization of single coil electromechanical RF relays, more specifically, HF3 and HF3 S series relays from TE Connectivity. The calibration procedure used to meet this goal should:
• be based on a low cost custom calibration kit ;
• be possible to execute in a short time frame;
• lead to accurate measurements.
1.3
Organization and Structure
This document is organized in six chapters, structured as follows:
Chapter 2 - Scattering Parameters
Introduces the concept of reflection and transmission coefficient, scattering parameters and N-port networks.
Chapter 3 - Vector Network Analyzer
Briefly describes the internal structure of VNAs and the measurement errors associated with them. It also presents the error term models used by the VNA and describes the Short-Open-Load-Thru (SOLT) and Thru-Reflect-Line (TRL) calibration techniques.
Chapter 4 - In-Fixture Custom Calibration Kit
Describes the procedure used to design and define the calibration standards for the cus-tom SOLT and TRL calibration kits. Presents improvements made over a preexisting custom SOLT calibration kit from TE Connectivity. It also introduces an experimental custom TRL kit used, among other things, to validate the quality of the Printed Circuit Board (PCB) production.
Chapter 5 - Frequency Characterization of RF Relays
Defines the figures of merit that describe the RF performance of electromechanical relays, details the calibration procedures and the measurement setup used, and reveals the results measured from sample RF relays from TE Connectivity.
Chapter 6 - Conclusions and Future Work
Concludes the overall dissertation, analyzing to what extent the objectives were met and suggesting improvements that can be made in a future work.
Chapter 2
Scattering Parameters
2.1
Introduction
Scattering Parameters (S-parameters) are used to describe the output(s) of electrical networks as a linear combination of its input(s). In this sense, S-parameters are similar to Y, Z and ABCD parameters. However, while S-parameters are defined as the relations of incident and reflected waves [6], the Y, Z and ABCD parameters are defined as relations of voltages and currents. S-parameters are measured using matched loads in contrast to the remaining parameters, which are measured using open or short circuits. At high frequencies matched loads are more suitable to use than open or short circuits, thus S-parameters are preferred when describing electrical networks operating at RF and microwave frequencies.
2.2
Reflection Coefficient
Reflection coefficient (Γ) is a parameter that quantifies how much energy of an incident wave, driven by a source of Z0 impedance, is reflected back from a load termination of different impedance ZL[7]. It can be determined by separately measuring the incident and reflected waves traveling on a transmission line.
Consider the transmission line depicted in figure 2.1, with length l, real Z0characteristic impedance and terminated with a load ZL 6= Z0. The voltage in the transmission line (V (d)) changes according to the distance from the load (d) and can be written as follows:
V (d) = Vinceγd+ Vrefe−γd (2.1)
where:
γ is the propagation constant of the transmission line; Vinc is the incident voltage wave;
Vref is the reflected voltage wave;
Figure 2.1: Transmission line with the source end at x = l and load end at x = 0 [8]
The total current standing wave (I(d)) can be obtained from equation 2.1 and it is defined as I(d) = Vinc Z0 eγd− Vref Z0 e −γd (2.2)
From equations 2.1 and 2.2 the load impedance at the line termination (d = 0) can be determined using the following equation:
V (0)
I(0)) = ZL = Z0
Vinc+ Vref Vinc− Vref
(2.3)
ΓL = Vref Vinc = ZL− Z0 ZL+ Z0 (2.4)
From ΓL it is possible to determine if:
• the load is passive, ZL > 0, when 0 6 |ΓL| 6 1 • the load is active, ZL < 0, when |ΓL| > 1
• the load is perfectly matched (ZL= Z0), when |ΓL| = 0
• the load is completely reflecting incident waves, when |ΓL| = 1
2.3
N-Port Networks
For the purpose of understanding how S-parameters describe the RF performance of a component, consider a two-port network similar to the one on figure 2.2.
S-parameters are defined in terms of input (a) waves and output (b) waves traveling waves and are defined as the square root of power waves (eqs. 2.5 and 2.6). As a result |a|2 and |b|2 can be respectively described as the incident and reflected power from the network.
Figure 2.2: Signal flow graph representation of the S-parameters of a two-port network
a = √Vinc Z0
b = √Vref Z0
- Output travelling wave (2.6)
A two-port network can be described by relating the reflected waves (b) with the incident waves (a) using the following set of equations:
b1 = S11a1 + S12a2 b2 = S21a1 + S22a2
(2.7)
Alternatively this set of equations can be written as the following matrix:
b1 b2 = S11 S12 S21 S22 a1 a2 (2.8)
Each S-parameter matrix defines an RF device at a single frequency and, therefore, several matrices are required to characterize a device over a range of frequencies.
Similar to the Z and Y parameters, S-parameters are obtained from the linear com-bination of the input and output variables of the network. S-parameters are however dimensionless, since both the input and output variables are defined as square root power waves (eqs. 2.5 and 2.6).
The parameters from the equations in 2.7 relate the input and output waves of a two-port network as follows:
S11= b1 a1 a2=0 S12= b1 a2 a1=0 S21= b2 a1 a2=0 S22= b2 a2 a1=0 (2.9)
Considering the two-port network from figure 2.2 and equations in 2.9, these parameters can be defined as follows:
termina-• S12 is the reverse gain of the network while port 1 is matched with a Z0 termination load.
• S21is the forward gain of the network while port 2 is matched with a Z0 termination load.
• S22 is the output reflection coefficient while port 1 is matched with a Z0 termination load.
By terminating port 1 or port 2 with a Z0 load, the respective port becomes matched and no reflection occurs.
The S-parameters of the network can also be described in terms of power:
• |S11|2 =
Power reflected from port 1 Power incident on port 1
• |S22|2 =
Power reflected from port 2 Power incident on port 2
• |S21|2 =
Power exiting from port 2
Power incident on port 1 = Power gain from port 1 to port 2
• |S12|2 =
Power exiting from port 1
Power incident on port 2 = Power gain from port 2 to port 1
2.4
Scattering Transfer Parameters
S-parameters are useful to describe the behavior of RF and microwave devices but it is difficult to use them when analyzing devices in chain. Consider obtaining the S-parameters of the network chain in figure 2.3. From what has been previously described, the natural procedure would be to represent the chain in figure 2.3 using signal flow graphs (such as the one on figure 2.2) and from there obtain the S-parameters of the total chain. This method may be doable for a chain with few elements, but becomes considerably laborious as the number of elements increases.
Figure 2.3: Cascaded two-port networks
A known alternative to this process is the usage of scattering transfer parameters, also known as T-parameters. Unlike the S-parameters, T-parameters can be defined as follows:
a1 = T11b2 + T12a2 b1 = T21b2 + T22a2
(2.10)
T-parameters are related to S-parameters by the equations on table 2.1 [10]:
Table 2.1: Conversion between S and T parameters
S11= T21 T11 S12 = T11T22− T12T21 T11 S21 = 1 T11 S22= −T12 T11 T11= 1 S21 T12 = −S22 S21 T21= S11 S21 T22= S12S21− S11S22 S21
Using the conversion equations in table 2.1 it is possible to transform the S-parameter matrices of the two-port networks in figure 2.3 in T-parameter matrices. This way, the T-parameter matrix of the total chain can be directly obtained from:
[Ttotal] = [TA][TB] (2.11)
Chapter 3
Vector Network Analyzer
3.1
Introduction
S-parameters are defined as ratios of input incident (a) waves and output scattered (b) waves. Measuring them requires a complex system, able to excite and measure both the incident and scattered waves at each separate port and over a wide range of frequencies. The first devices with such capabilities were introduced in the 60s in the form of a Vector Network Analyzer (VNA), but it was only around the mid 80s that the first VNAs with integrated microprocessors became available. The 8753A from Hewlett-Packard (HP) was the first fully error-corrected RF network analyzer and, because of its low cost and high capability, it quickly became the industry standard [11]. VNAs such as the 8753A intro-duced several capabilities that greatly improved the RF design of that time, becoming the basis of forthcoming VNAs.
In this chapter, a brief description of the internal architecture of a VNA will be ex-plained, followed by an overview of the VNA measurement errors and respective vector error correction.
3.2
Internal Block Diagram
The first measurement systems could only measure the transmission or reflection re-sponse of a Device Under Test (DUT) each direction at a time. These devices consist of three receivers and a directional device. Three receivers to respectively measure the incident, reflected and transmitted waves, and a directional device to isolate the reflected wave from the incident wave.
Measuring the S-parameters of a two-port network with these systems requires the user to manually change the port connections to the DUT. By connecting port 1 to the input and port 2 to the output of the DUT, the user is able to perform a forward measurement, thus obtaining S11and S21. To obtain the remaining parameters port 1 has to be connected to the output of the DUT and port 2 to the input.
Modern VNAs do not have this limitation, since there are several key components responsible for switching the RF path of the measurement setup. These components enable the VNA to automatically split and measure incident and reflected waves at all ports, without the need to manually change the measurement setup (figs. 3.1 and 3.2).
Figure 3.2: Simplified test set block diagram of the E8361 VNA used in this work [5]
The key components in a VNA are [13]:
RF or microwave source: Generates the signal that is used to stimulate the DUT. Some of the important characteristics of an RF source are the frequency range, power range, sweep speed, frequency stability and harmonic and spurious content. Modern VNAs may have up to one source per port.
RF test set: Switches and separates the incident and reflected waves, enabling the VNA to measure them separately. Modern VNAs come with integrated RF test sets. There are however cases where it is necessary to use an external RF test set, for instance when performing high power measurements.
Receivers: Measure both the magnitude and phase of the signal. In order to deter-mine the phase, all receivers must share the same local oscillator. Usually the signals are down converted and measured at a lower Intermediate Frequency (IF).
CPU: Modern VNAs contain embedded personal computers that support operating systems such as Microsoft Windows. This way modern VNAs take advantage of the cur-rently existing programming environments to create custom firmware, improving the per-formance of the system beyond the limitations of the hardware.
Front Panel: Includes a digital display and several keys to access different functions and menus. As VNAs are versatile measuring instruments packed with different function-alities, the user interface tends to be a bit complex.
Rear Panel: Is the part of the VNA where most input/output interfaces are located. This may include: video display outputs, USB interfaces, LAN interfaces, ports to attach external voltage sources, voltmeters, GPIO buses and pulse generators.
3.3
Measurement Errors
Just like any other measurement equipment VNAs are prone to different types of errors. In VNAs there are three different types of measurement errors [14], [15].
• Random errors are unpredictable and, therefore, cannot be removed. These errors are mainly caused by instrument noise and the repeatability of switches, cables and connec-tors. Even though random errors cannot be removed, they may be reduced by narrowing the IF bandwidth of the VNA and by averaging several measurements [16].
• Systematic errors are caused by imperfections in the test equipment and overall test setup. These errors are stable and repeatable, thus predictable. They can be char-acterized and removed mathematically from the measurement system through a process known as VNA calibration.
• Drift errors occur when the performance of the equipment changes after the sys-tematic errors have been mathematically removed. It is mainly caused by temperature fluctuations in the measurement environment and as such can be minimized by using a temperatucontrolled environment. Drift errors can be removed by mathematically re-moving the systematic errors once again [17].
Consider the depiction in figure 3.3 of the forward measurement of a DUT. Where ’R’ is the reference receiver that measures the incident wave coming from the RF source and into the DUT. The receiver ’A’ measures the wave reflected from the DUT and the receiver ’B’ measures the wave transmitted through the DUT.
Figure 3.3: Systematic measurement errors [14]
The systematic errors identified from this three-receiver measurement system can be described as:
Directivity Error:
All VNAs require directional couplers or bridges to separate incident and reflected waves. With an ideal coupler, the receiver ’A’ would only measure the wave reflected from the DUT. In practice, part of the incident wave leaks to the receiver ’A’. Leakage paths such as these contribute to the directivity error.
Source Match Error:
’A’. In reality, the wave reflected from the DUT suffers multiple reflections between the source port and the DUT itself. These reflections are also measured by the receiver ’A’, resulting in a source match error.
Frequency Response Reflection Tracking Error:
Reflection measurements result from the ratio between the reflected wave in receiver ’A’ and the incident wave in receiver ’R’ (A/R). On an ideal system the frequency response of both receivers would be identical. However, variations between the reference and test signal paths causes magnitude and phase to change with frequency, resulting in a frequency response reflection tracking error.
Isolation Error:
Ideally, the receiver ’B’ measures only the traveling wave transmitted through the DUT. However, part of the incident wave is able to leak through different paths and be measured by the receiver ’B’. This leakage is known as crosstalk and is the cause for isolation error.
Load Match Error:
During a transmission measurement, the receiver ’B’ should measure the total traveling wave transmitted through the DUT. In practice, some of the transmitted wave is reflected at the load port back into the DUT. As a result, the receiver ’B’ measures only a fraction of the transmitted wave, causing load match error.
Frequency Response Transmission Tracking Error:
Transmission measurements result from the ratio between the transmitted wave in re-ceiver ’B’ and the incident wave in rere-ceiver ’R’ (B/R). On an ideal system the frequency response of both receivers would be identical. However, variations between the reference and test signal paths causes magnitude and phase to change with frequency, resulting in a frequency response transmission tracking error.
3.4
Vector Error Correction
One of the strong advantages of a VNA is its versatility, since it can be used with great precision in a wide variety of configurations. However, the VNA needs to be calibrated for each of these configurations. The VNA owes its measurement accuracy to the correction of the magnitude and phase responses, through a process commonly referred to as vector error correction or simply VNA calibration. A VNA calibration, similarly to the calibration of any electronic measurement equipment, checks if the accuracy of the measurements is within specification and performs any necessary corrections.
Electronic measurement equipment is commonly calibrated by the manufacturer once a year and most of the times using certified measurement equipment of higher quality. Nevertheless, modern VNAs are also able to characterize their hardware on their own. This allows them to compensate for its internal effects, as well as the effects of connectors, cables, test-fixtures and probes. This procedure can be easily performed by the user on site and allows for a far more accurate measurement of the DUT.
The VNA calibration is achieved by measuring well known calibration standards, which the VNA uses to determine the systematic errors. Once the systematic errors are de-termined, the VNA mathematically corrects the raw measurement and displays a more accurate measurement of the DUT.
By calibrating the VNA, the user is not improving the actual performance of the equip-ment. With this procedure the user is in fact characterizing and compensating the limi-tations of the hardware. The correction of the systematic errors is also performed by the manufacturer before shipping, so that measurements taken directly from the ports of the VNA are displayed correctly. The factory calibration is often saved in the memory of the VNA and applied once the software is initialized.
3.4.1
Twelve-Term Error Model
In the process of characterizing the systematic errors the calibration algorithm creates a model representation of the measurement setup that represents the systematic errors and the DUT separately. One of the error models that is commonly used on VNAs is the twelve-term error model (fig. 3.4), which is based on the simultaneous measurement of three receivers. The twelve-term error model is in fact made up of two separate six-term error models (figs. 3.4a and 3.4b).
(a) Forward error model
(b) Reverse error model
In order to determine the twelve error terms in table 3.1 twelve equations are required. As a result, one measurement must be made for each of the error terms. This does not imply that one error term can be determined from a single measurement. Since the forward model on figure 3.4a and the reverse model on figure 3.4b are independent, six independent measurements must be made for each direction.
The leakage terms are determined from the isolation calibration. However, since the crosstalk on modern VNAs is typically below the noise floor of the measurement system, the isolation calibration is commonly omitted [13].
Table 3.1: Systematic error terms
EDF Forward Directivity EDR Reverse Directivity
EM1F Forward Port 1 Match EM1R Reverse Port 1 Match
ERF Forward Reflection Tracking ERR Reverse Reflection Tracking
ETF Forward Transmission Tracking ETR Reverse Transmission Tracking
EM2F Forward Port 2 Match EM2R Reverse Port 2 Match
ELF Forward Leakage ELR Reverse Leakage
To differentiate the raw measurements from the actual measurements of the calibration standards, the notation presented on table 3.2 is used.
Table 3.2: Raw parameters measured from the calibration standards
Short Open Load Thru
b1M a1M S Short 11M S Open 11M S11MLoad S11MT hru b1M a2M —– —– S Load 12M S12MT hru b2M a1M —– —– S Load 21M S21MT hru b2M a2M S Short 22M S Open 22M S Load 22M S22MT hru
SOLT Calibration
The calibration method most commonly used to determine the systematic errors of the VNA is the SOLT calibration. This method consists in measuring three one-port calibration standards (an open, short and load) on each port and a thru standard connected to both ports [19]. Additionally, if the isolation calibration is performed, another measurement is taken with each port terminated with a load standard. During the calibration procedure a total of 10 or 12 measurements are taken, depending if the isolation calibration is omitted or not.
Table 3.3: S-parameter matrices of ideal calibration standards
Short Open Load Thru
−1 0 0 −1 1 0 0 1 0 0 0 0 0 1 1 0
For a SOLT calibration using ideal standards (tbl. 3.3) the six error terms calculated from the measurements of one-port calibration standards would be:
EDF = S11MLoad, EDR = S22MLoad
EM1F =
S11MOpen+ S11MShort− 2EDF S11MOpen− SShort
11M
, EM2R =
S22MOpen+ S22MShort− 2EDR S22MOpen− SShort 22M ERF = −2(S Open 11M − EDF )(S Short 11M − EDF ) S11MOpen− SShort 11M , ERR = −2(S Open 22M − ERF )(S Short 22M − ERF ) S22MOpen− SShort 22M (3.1)
ELF = S21MLoad, ELR = S12MLoad
(3.2)
and the four terms from the measurement of the thru standard would be:
EM1R =
ST hru
22M − EDR
[ERR + EM2R(S22MT hru− EDR)]
, EM2F =
ST hru
11M − EDF
[ERF + EM1F (S11MT hru− EDF )]
ET F = (S21MT hru− ELF )(1 − EM1F · EM2F ), ET R = (S12MT hru− ELR)(1 − EM2R · EM1R)
(3.3)
In reality, there are no ideal calibration standards. SOLT calibration standards may suffer from attenuation or open-end fringing capacitance. To overcome these errors, the SOLT standards are characterized and a model representation that can be interpreted by the VNA is created. The accuracy of the SOLT calibration relies on the definition of the models for the calibration standards. As a result, any differences between the models and the actual calibration standards will be considered as systematic errors and the VNA will erroneously compensate for them.
SOLT calibration is commonly used by RF engineers to reduce the measurement errors and set the measurement reference plane at the coaxial connection between the DUT and the measurement setup. This calibration procedure is very useful when measuring connectorized devices such as amplifiers or filters. To perform this calibration technique commercial grade SOLT calibration kits are often used. Some of which are produced by manufactureres certified by the National Institute of Standards and Technology (NIST) and other international standards organization members [20].
Most calibration kits such as the one on figure 3.5, lack the model definition for non-zero length thru standards. As a result, SOLT calibration can only be performed when measuring insertable devices (fig. 3.6), since it is only under these circumstances that the zero length thru standard can be measured. In contrast, when measuring non-insertable
devices the Short-Open-Load-Reciprocal (SOLR) calibration method [21] is preferred over the SOLT calibration, as it provides accurate results without the need to completely define the thru standard.
Figure 3.5: 85052D 3.5 mm SOLT calibration kit from Agilent Technologies
Figure 3.6: Classes of DUT [22]
There are often cases when it is necessary to measure non-connectorized devices, where commercially available calibration kits cannot be used or are simply too expensive. In such cases, it is possible to design and produce custom calibration kits. Custom SOLT calibration standards are considerably difficult to create on planar transmission media with precision [23]. In those cases, alternative calibration methods, such as TRL, can be
3.4.2
Eight-Term Error Model
The eight-term error model (fig. 3.7) is based on the simultaneous measurement of four receivers (fig. 3.2). Both the eight and twelve-term error models represent the same system and are therefore interchangeable [24]. This makes it possible to use the eight-term error model on a VNA with three receivers, and the twelve-term error model on a VNA with four receivers.
Figure 3.7: Two-port eight-term error model [18]
Analyzing the eight-term error model on figure 3.7 as a chain of two-port networks, the transfer matrix of the total chain becomes:
TM = TXTDU TTY (3.4) or TM = 1 X21Y21 1 −X22 X11 −∆X TDU T 1 −Y22 Y11 −∆Y (3.5)
where ∆ is the determinant of the matrix.
From equation 3.5 it is possible to identify three error terms for port 1 (−X22, X11 and −∆X), three error terms for port 2 (−Y22, Y11 and −∆Y), and one transmission error term (X21Y21).
These seven error terms can be calculate by measuring three two-port calibration stan-dards.
From those measurements twelve independent equations are generated, from which only seven are used to calculate the error terms. With this redundancy, it is possible to use partially unknown calibration standards.
TRL Calibration
Since the twelve and eight term error models are interchangeable, the SOLT calibration can also be used to determine the systematic error terms from the eight-term error model [25]. However, the TRL calibration [26] is often preferred, since it is able to take advantage of the redundancy of the eight-term model. It does so by using only one completely known calibration standard and two other partially known standards. The TRL calibration procedure consists in measuring a thru connection, a reflect and a line standard (fig. 3.8). The thru calibration standard is created using a zero length thru and as such it can be easily defined, since S11 = S22 = 0 and S21 = S12 = 1. By terminating both ports with the same reflective standard, a two-port calibration standard with S11 = S22 is obtained. Finally, the line standard is used to set the reference characteristic impedance (Z0) and, as such, only the match of this standard is of interest (S12 and S21 can be unknown).
Figure 3.8: TRL calibration kit with two delay lines [27]
require all calibration standards to be completely known. However, for the TRL calibration to be accurate, the calibration standards must fit the following set of requirements [28]:
Zero Length Thru
• Must have no loss;
• S21= S12= 1∠0◦;
• S11= S22= 1.
Non-Zero Length Thru
• Z0 of the thru must be the same of the line;
• If the thru is used to set the reference plane, the insertion phase or electrical length must be specified. If a non-zero length thru is specified to have zero delay, the reference plane is set in the middle of the thru.
Reflect
• The reflection coefficient (Γ) must be identical on both ports;
• If the reflect is used to set the reference plane, the phase response must be well-known and specified.
• |Γ| should ideally be equal to 1;
Line
• Z0 of the line sets the reference impedance of the measurement (i.e., S11 = S22= 0).
• The difference in insertion phase between the thru and line must be between 20◦ and 160◦;
• Optimal line length is 1
4 wavelength of the insertion phase relative to the thru at the middle of the desired frequency span;
On a TRL calibration, the parameter that sets the reference impedance is the char-acteristic impedance (Z0) of the line standard. Since Z0 can be precisely known from the physical dimensions of the line standard, the TRL calibration is often preferred on metrology grade calibrations [29].
Despite being potentially the most accurate method of VNA calibration [30], there are some limitations for the TRL calibration technique. The main disadvantage of the TRL calibration is the limited bandwidth of the line standard. This limitation requires the production of several line standards to extend the bandwidth of the calibration. However, for calibrations at lower frequencies, the dimensions of the line standard may become unpractical. To avoid this problem, the line standard can be replaced by a load standard to perform what is known as the Thru-Reflect-Match (TRM) calibration [31].
3.5
VNA Models for the Calibration Standards
Calibration standards are far from ideal, therefore, VNAs store a model representation of each standard [32]. Differences between the standard and respective model may lead to inaccurate corrections, as they are eventually considered part of the error terms. With careless or intensive use, the calibration standards eventually become degraded and no longer correspond to the respective VNA model. The higher the quality of the calibration kit, the more mechanically robust it will be and the better will the models correspond to the actual calibration standards.
Open Standard
An open standard suffers from fringing capacitance, which adds phase shift to the standard. To account for these effects, the open standard is described with the model in
C(f ) = C0+ C1f + C2f2+ C3f3 (3.6)
Figure 3.9: Model of an open standard
Short Standard
The short standard is typically described with the model in figure 3.10 and defined using an inductance polynomial dependent of frequency (eq. 3.7).
L(f ) = L0+ L1f + L2f2+ L3f3 (3.7)
Load Standard
The load standard is, mainly, responsible for the correction of the directivity error terms [23]. The most common model for this standard includes only a resistance and a delay line (fig. 3.11). The value of load impedance is typically fixed to be 50Ω, but can also be set to any arbitrary value.
Figure 3.11: Model of a load standard
Thru/Line Standard
A zero length thru, a non-zero length thru and a line standard are all defined by the same model (fig. 3.12), but with different parameters. The zero length thru is the easiest to model, since it has no attenuation, characteristic impedance or delay. The second easiest is the line standard, since it only requires the delay to be specified. Finally, the hardest is the non-zero length thru, which requires all parameters to be specified.
Chapter 4
In-Fixture Custom Calibration Kit
4.1
Introduction
With the miniaturization of RF components, the need to characterize connector-less devices becomes a necessary step in the design of further integrated RF systems. The interface between a connector-less DUT and the VNA is achieved by connecting the DUT to a test-fixture, that in turn can be connected to the VNA. A test-fixture can only be used to measure devices with the footprint for wich it was designed. Ideally, a test-fixture would connect the measurement setup to the DUT without adding any losses or mismatch errors, thus allowing the VNA to measure the DUT accurately without the need for calibration. Unfortunately, test-fixtures add errors to the measurements of the DUT and, in order to reduce them, in-fixture calibration standards are used [33]. In-fixture calibration standards allow the VNA to identify the characteristics of the test-fixture and reduce measurement errors through vector error correction. In the end, the accuracy of the measurements depends directly on the quality of the standards and similarity with the test-fixture.
The first part of this chapter introduces the RF relays from TE Connectivity as the DUT used in this work. This is followed with a detailed explanation of the design and im-plementation of two custom calibration kits, one SOLT and one TRL, and their respective test-fixtures. Improvements made over a preexisting kit from TE Connectivity are also
covered. As well as, the design of an experimental custom TRL calibration kit that was used, among other things, to validate the manufacturing process.
4.2
RF Relays from TE Connectivity
For the purpose of this work two miniaturized single coil RF relays from different series were selected:
• The HF3 53 relay on figure 4.1a is part of the HF3 series. Relays from the HF3 series offer good RF characteristics up to 3 GHz. They are also available for both 50 Ω and 75 Ω systems and are able to carry a maximum of 50 W at 2.5 GHz (see appendix A).
• The HF3 55 S relay from figure 4.1b is part of the HF3 S series (where the ’S’ stands for Shield). Relays from the HF3 S series offer improved performance over the HF3 series relays. Just like the HF3 series relays, they are also rated up to 3 GHz and are also available for both 50 Ω and 75 Ω systems, however, they can carry a maximum of 100 W continuous RF power at 3 GHz (see appendix B).
(a) HF3 53 (b) HF3 55 S
Figure 4.1: RF relays under test
Despite both series having different RF performance, they share the same package dimensions. HF series relays can be either single coil monostable, single coil bistable or
For the purpose of this work, only 50 Ω single coil HF relays will be considered.
Monostable Relays
Monostable relays (fig. 4.2a) have a spring that forces them to return to a known state as soon as the coil is deactivated [34]. The spring in HF3 and HF3 S monostable relays forces them to close the Normally-Closed (NC) contact. For that reason, HF series monostable relays can only keep the Normally-Open (NO) contact closed for as long as the coil is active, thus continuously consuming power.
Bistable Relays
Bistable relays (fig. 4.2b) do not return to a known state when the coil is deactivated. The relay is switched by inverting the polarity of the voltage that was previously applied to the coil. When a contact is closed, it will remain closed, even if the coil is deactivated, and will only open when the coil is activated with reverse polarity. This is a great advantage, since bistable relays only consume power when switching between states.
(a) Monostable HF relay (b) Bistable HF relay
4.2.1
Internal Structure of the HF Series Relays
HF series relays are composed of several parts, each with a specific function (fig. 4.3).
Figure 4.3: Exploded view of an HF3 relay [2]
• Metal Cover - Is only present in HF3 S relays. It goes over the plastic cover and allows for a better RF shielding from the neighboring circuits.
• Plastic Cover - Covers the other remaining components of the relay, maintaining them hermetically sealed. It prevents dust and other particles from compromising the RF performance or even damaging the relay.
• Coil - When activated by an electric current, generates a magnetic field to move the armature.
• Yoke - Increases the magnetomotive force applied to the armature.
• Spring - It is only present in monostable relays. It forces the relay to change back to the initial state when the coil is deactivated.
• Armature - It is the mobile component of the relay. It has a ferromagnetic piece attached that moves the armature as the coil is activated.
to-• RF Carrier - It is the base of the relay. It carries the RF signal from the common terminal across the RF switching contact to either the NO or the NC terminal (the path is bidirectional) (fig. 4.4).
The HF3 and HF3 S are Single Pole Double Throw (SPDT) relays based, on an RF carrier designed using a Y shaped co-planar structure. The design of the carrier determines the characteristic impedance of the relay. The 50 Ω relay, in comparison with the 75 Ω, has a smaller gap between the signal lines and the grounding (fig. 4.4a). It also has bifurcated RF switching contacts, different from the single straight structure used on the 75 Ω relays (fig. 4.4b).
(a) 50Ω version (b) 75Ω version
Figure 4.4: Design principle of the HF3 series relays [2]
The trigger system of the HF relays is based on a standard magnetic circuit that is polarized according to the monostable or bistable switching characteristics. HF series relays have a total of 22 terminals: 3 to carry the RF signal, 2 or 4 to trigger the coil (depending if the relay is monostable or bistable) and the remaining terminals must be grounded to provide optimal shielding (see appendices A and B).
4.3
Calibration Standards Design and Definition
The design of the DUT determines the design of the test-fixture which, in turn, de-termines the design of the calibration kit. In order to connect the DUT to the VNA, a test-fixture, similar to the one on figure 4.5, was designed.
Figure 4.5: Coaxial test-fixture [35]
This design includes an end launch SubMiniature Version A (SMA) connector, to change the propagation media from coaxial to Co-Planar Waveguide with Lower Ground Plane (CPWG), and a transmission line of Z0reference impedance, to connect the SMA connector to the DUT. Test-fixtures however, introduce measurement errors due to discontinuities in the successive transitions [36]. As previously covered in section 3.4, these errors can be reduced through VNA calibration. In order for the calibration to set the reference plane at the pads of the DUT, the measured calibration standards must be defined at the exact same position where the DUT will be placed. As such, the length of the transmission lines used on the calibration standards must be the exact same length as the ones used on the test-fixture (figs. 4.6, 4.7 and 4.8).
Figure 4.6: Custom SOLT calibration kit
Figure 4.8: Measurement reference plane set by the calibration kits
The in-fixture calibration standards were designed using Advanced Design System (ADS), an Electronic Design Automation (EDA) tool from Keysight Technologies. The Z0 transmission lines on the calibration standards and test-fixture where designed using CPWG technology (fig. 4.9), in order to be compatible with the RF carrier of the HF series relays (fig. 4.4).
Figure 4.9: Cross section of a CPWG transmission line [37]
For the purpose of this work, two custom in-fixture calibration kits were designed to characterize an HF3 53 and HF3 55 S relay. The SOLT and TRL custom calibration kits were designed on the same Printed Circuit Board (PCB) to ensure that the manufacturing
different calibration kits it was possible to compare the different implementations and determine which of them is most suitable for in-fixture measurements.
The custom calibration kit and respective test-fixture were manufactured using a lam-inate of RO4350B substrate (see appendix C), from Rogers Corporation, with 0.762 mm of thickness. Due to the thin thickness of the substrate, the PCBs with the calibration kits and test-fixture where fixed on aluminum boards to avoid bending or even breaking them (fig. 4.10). The aluminum boards were designed in the Institute of Telecommu-nications (IT) (see appendix D) and manufactured at the Department of Physics of the University of Aveiro.
Figure 4.10: Aluminum board for Kit IT v2 (on the left) and respective test-fixture (on the right)
4.3.1
Short Standard
The short standard was based on the design in figure 4.11. This standard was created by shorting the transmission line with the top and bottom ground planes.
Figure 4.11: Design of a CPW short circuit [38]
The bottom ground plane was connected to the transmission line using a plated through hole (figs. 4.12 and 4.13).
Figure 4.13: In-fixture short standard from Kit IT v2
The short standard has |Γ| = 1, 180◦ of phase shift and can be used to define the mea-surement reference plane [39]. After calibrating using the 85052D kit, the meamea-surement reference plane is set at the SMA connector. Thus, the VNA will measure the reflection coefficient (Γ) of everything that follows the SMA connector, i.e., the transmission line together with the short circuit at the end of the line. It is important to determine accu-rately where the measurement reference plane is set, as it might be needed to eventually compensate the offset of the remaining standards. The offset of the measurement reference plane imposed by the short standard can be determined using the port extension tool [40], available on the VNA software.
The port extension tool mathematically shifts the measurement reference plane, from the connector to the short standard, by the amount of delay set by the user. To determine the delay between the reference plane set at the SMA connectors and the short circuit termination, the port extension tool must be shifted until the phase measured by the VNA reaches 180◦. This delay should be stored, since it is useful to determine offset delays between the remaining calibration standards and the short standard. When the short standard is used to set the reference measurement plane, it is unnecessary to define the offset loss, delay and impedance.
Since the short standard is also the reflect standard of the TRL calibration kit, they share the same standard definition.
4.3.2
Open Standard
The open standard was based on the design in figure 4.14. This standard was created by ending the transmission line on an open termination (figs. 4.15 and 4.16).
Figure 4.14: Design of a CPW open circuit [38]
Figure 4.16: In-fixture open standard from Kit IT v2
The open standard is highly reflective, with a |Γ| = 1 and a theoretical phase shift of 0◦. In practice the open standard typically has some phase shift due to fringing capacitance.
To determine the fringing capacitance, the port extension tool is used to shift the measurement reference plane to the same position the short standard is defined at. If at this point the VNA displays the reflection coefficient shifting anti-clockwise (fig. 4.17), the open standard has an offset delay. To measure the offset delay, the port extension should be shifted until the phase response of the open standard is monotonically negative (fig. 4.18b). The offset delay of the open standard is the difference between the delay extracted from the short standard, using the port extension tool, and the delay displayed on the open standard with the phase monotonically negative [39]. As soon as the reference measurement plane is set, the fringing capacitance can be directly measured from the VNA.
(a) without port extension
(b) with port extension shifted to the position of the short standard
4.3.3
Load Standard
The load standard uses two 100Ω CH0603-100RJN Surface Mount Technology (SMT) resistors, from Vishay (see appendix E), the end of the transmission line (figs. 4.19 and 4.20). By using two 100Ω resistors in parallel instead of only one 50Ω resistor, it is possible to reduce the parasitic inductance by half [28].
Figure 4.19: Design of the load standard from Kit IT v2
Figure 4.20: In-fixture load standard from Kit IT v2
Typically, the offset delay and impedance of the load standard are obtained using time domain gating [39]. Time domain gating is a feature available on some VNAs that is used to remove the effects from the test-fixture. Unfortunately, it was not possible to use the
used. To compensate for this shortcoming, special care was taken in the design of the load standard (figs. 4.19 and 4.20), so that the pads of the resistors would be located at the same length as the ground via realizing the short standard (figs. 4.12 and 4.13). This guarantees that the reference measurement plane is defined at the SMT resistors and, as a result, it is unnecessary to define the offset loss, delay and impedance.
4.3.4
Thru Standard
The thru standard is achieved using a transmission line (figs. 4.21 and 4.22) with double the length of the transmission lines used on the previous standards. As a result, the refer-ence measurement plane was set at the middle of the transmission line. Since a zero length thru is designed, there is no need to characterize the offset loss, delay or impedance for its VNA model. The transmission lines on the test-fixture and every calibration standard were designed to share the exact same characteristic impedance.
Figure 4.22: In-fixture thru standard from Kit IT v2
4.3.5
Line Standard
The length of the line standard defines the frequency range of the calibration and it must be inserted into the VNA in the form of a delay.
One of the requirements for the line standard states that the difference in electrical length between the line and thru must be within 20◦and 160◦. In practice, this requirement limits the bandwidth of the line standard according to the following condition:
fmax fmin
≤ 8 (4.1)
where:
fmin is the minimum frequency supported by the line; fmax is the maximum frequency supported by the line.
The frequency range of the TRL calibration kit was set to start at 100 MHz, so that the line standard would not be too long, and stop at 3 GHz. However, because of the condition 4.1, it is not possible to design a single line that is valid for the desired frequency range. To overcome this, the frequency range of the calibration was split over two lines at the optimal break frequency (eq. 4.2) [35].
Therefore, two lines were designed to cover different frequency ranges (figs. 4.23 and 4.24):
• Line 1 with a frequency range from 100 MHz to 600 MHz (fig. 4.25a);
• Line 2 with a frequency range from 600 MHz to 3 GHz (fig. 4.25b).
Once the bandwidth of each line is defined, equation 4.3 [28] is used to determine the length of the line.
Electrical Length (cm) = 15000 × V F
fmin(M Hz) + fmax(M Hz)
(4.3)
Where the velocity factor, VF = 1 p(ef f)
.
Using equation 4.3 length of line 1 and line 2 was calculated to be respectively 13.00 cm and 2.52 cmm.
Figure 4.23: Design of the line standard from Kit IT v2
(a) Line 1
(b) Line 2
4.4
Improvements over the V032K Calibration Kit
Design
The design of the custom SOLT calibration kit (fig. 4.26b) was based on the design of a preexisting kit from TE Connectivity, the V032K SOLT calibration kit (fig. 4.26a). Some calibration standards that could be improved were identified by taking a closer look at the design of the V032K calibration kit.
(a) V032K (b) Kit IT v2
Figure 4.26: Custom SOLT calibration kit from TE Connectivity (a) and IT (b)
The V032K calibration kit is based on the 4350B substrate from Rogers Corporation (see appendix C). Using the LineCalc tool from ADS it was possible to determine the characteristic impedance (Z0) of the transmission lines used on this kit, which (based on their dimensions and substrate properties) resulted in Z0 = 52Ω.
A closer look at the design of the open standard reveals that the transmission line ends on a rounded shape (fig. 4.27a). As seen in [38], a flat termination is a more appropriate end of line, as the reflection plane is better defined (fig. 4.27b).
(a) V032K
(b) Kit IT v2
Figure 4.27: Improvements over the design of the open standard
A closer look at the design of the short standard reveals a transmission line ended on a plated ground via that connects only to the bottom ground plane and leaving a round shaped gap with the top ground planes (fig. 4.28a). As seen in [38], a flat termination connecting both the top and bottom ground planes is a more appropriate end of line, as the reflection plane is better defined (fig. 4.28b).
(a) V032K
(b) Kit IT v2
Figure 4.28: Improvements over the design of the short standard
The design of the Kit IT v2 also corrects the dimensions of the transmission lines on the calibration standards and test-fixture so that Z0 becomes 50Ω. This new design introduces these improvements over the design of the V032K calibration kit.
4.5
Experimental custom TRL Calibration Kit
The first calibration kit to be manufactured, was an experimental TRL calibration kit designed based on [35]. This kit was used to validate the manufacturing process, before or-dering the production of the Kit IT v2. This experimental custom TRL calibration kit, also referred throughout this document as Kit CGH35015F, was used to characterize a Gallium
Nitride (GaN) transistor from Cree, the CGH35015F (see appendix F). The results ob-tained from the characterization of this transistor were compared with the results obob-tained from its respective simulation model, which was provided by the manufacturer. From this comparison it was possible to validate the design and production of Kit CGH35015F.
Kit CGH35015F was designed to calibrate the VNA between 500 MHz and 3 GHZ, thus avoiding the use of longer and unpractical line standards. The PCBs with the Kit CGH35015F and respective test-fixture (fig: 4.30) were also fixed in aluminum boards (see appendix D). The design of the kit and test-fixture uses microstrip transmission lines (fig. 4.31) to match with the package of the transistor (see appendix F).
Figure 4.29: Custom TRL CGH35015F calibration kit
Figure 4.31: Cross section of a microstrip transmission line [37]
Once the calibration kit and test-fixture were manufactured (see appendix G), the S-parameters of a CGH35015F were measured using the setup in figure 4.32. The attenuator, connected on port 2, was used to prevent damaging the VNA in case the transistor started to oscillate. To compensate the effect of the attenuator on the measurements, the VNA was set with a lower IF bandwidth and averaging was turned on, in order to increase the dynamic range of the system [16].
Those measurements were afterwards compared with the ADS simulation using the model of the transistor, provided by the manufacturer (fig. 4.33).
Figure 4.33: Simulation setup for a CGH35015F GaN transistor
The S-parameter measurements were compared with the transistor biased at different points (figs. 4.34, 4.35 and 4.36):
From the comparison in figures 4.34, 4.35 and 4.36, it can be verified that the S-parameters measured are similar to the S-S-parameters that resulted from the simulation, thus validating the design procedure and production of the TRL calibration kit.
The TRL calibration kit results from the combination of three calibration standards: the reflect, the thru and the line. Since both the SOLT and TRL calibration kits are designed on the Kit IT v2 PCB, some of the calibration standards are shared between both kits. This is the case for the short and thru standards. Only the line standard had to be exclusively designed for the TRL calibration kit.
With Kit IT v2 manufactured (see appendix H), it became possible to reduce the errors of the test-fixture from the measurements, by performing either SOLR or TRL calibration.
4.6
Cost Evaluation
Table 4.1 compares the production cost for the V032K and Kit IT v2 calibration kits and respective test-fixtures. All values were obtained by requesting quotes directly from the manufacturers or resellers and include shipping costs when applied.
Table 4.1: Comparison of production costs
Material Description Kit V032K Kit IT v2
PCB 529.60e 65e
Aluminum Board ——— 45e
SMA Connectors 130.75e 93.60e
SMT Resistors 9.7e 9.5e
Total 640.45e 213.10e
the Kit IT v2 was ordered to a local manufacturer with little experience in working with the 4350B substrate. On the other hand, the PCBs of the V032K calibration kit were produced by a Swiss manufacturer with years of experience in manufacturing boards for RF applications. Even though the PCBs for Kit IT v2 required aluminum boards it was cheaper to produce them than it was to produce the PCBs for kit V032K.
The cost of the SMA connectors used on Kit IT v2 is different from the cost of the connectors used on Kit V032K. That is because, while the price of the connectors used on Kit IT v2 was obtained from an European reseller, the price of the connectors used on kit V032K were obtained directly from the South Korean manufacturer. Therefore, shipping costs, customs duties and taxes had to be considered.
No known reseller had the SMT resistors used on the V032K kit available at the time of this work. As a result, the quotation for these resistors was requested directly from the manufacturer.
