High-Performance Isolated
Bidirectional DC-DC Converter
Nuno Martins Alves
Mestrado Integrado em Engenharia Eletrotécnica e de Computadores Supervisor: Prof. Adriano da Silva Carvalho
Company’s Supervisor: Eng. José Pedro Fortuna Araújo - AddVolt
Nos últimos tempos, fatores como sensibilização social e pressões políticas têm levado ao alterar da maneira como a sustentabilidade ambiental é encarada, resultando no constante impulsionar e adotar de energias renováveis assim como de novas maneiras de nos transportarmos. Em am-bas estas vertentes, a eletrónica de potência assume um papel crucial e determinante para o seu desenvolvimento; em particular, conversores DC-DC bidireccionais isolados podem ser encontra-dos num vasto leque de aplicações, desde sistemas de produção eólica ligaencontra-dos à rede nacional até estações de carregamento de veículos elétricos.
Partindo desta contextualização, esta dissertação, proposta pela AddVolt, visa o desenvolvi-mento de um conversor DC-DC bidireccional isolado de média potência para integração numa aplicação de travagem regenerativa num camião frigorífico. Este documento surge assim como um compêndio da solução que visa dar resposta à necessidade sentida, cobrindo todas as diversas etapas desde o levantamento de requisitos até à validação de um protótipo experimental.
Como resultado do levantamento de literatura, a topologia Dual Active Bridge é analiticamente explorada e caracterizada. Um adicional nível de liberdade foi tido em conta através do desenho do transformador de alta frequência que assegurará o isolamento galvânico. Uma série de simulações computacionais são apresentadas, onde o funcionamento do conversor em regime estacionário pode ser estudado. Uma malha de controlo de tensão de saída é dimensionada e testada para diversos cenários de carga.
A assemblagem experimental do conversor DAB é descrita e, para além de delineados os principais aspetos inerentes a software e hardware, enfâse é dado à implementação de dois trans-formadores baseados em designs previamente apresentados. Por fim, ensaios práticos são levados a cabo, sendo que prova de conceito é garantida através de um protótipo experimental a operar na ordem dos 2 kW.
Por fim, propostas de trabalho futuro são recomendadas para que esforços suplementares pos-sam levar ao alcançar de ainda mais promissores resultados.
Lately, aspects as social awareness and political pressures have led to a shift in the way environ-mental sustainability is faced, resulting in a constant drive and adoption of renewable energies as well as clean ways of transportation. In both these strands, power electronics takes a crucial and decisive role in its development; in particular, bidirectional isolated DC-DC converters can be found in a broad spectrum of applications, ranging from grid-connected wind systems to electric vehicles’ charging stations.
On this basis, this dissertation, commissioned by AddVolt, aims the development of a medium-power bidirectional isolated DC-DC converter for integration in a regenerative breaking applica-tion for a refrigerator truck. This document emerges as a compendium of the proposed soluapplica-tion for the perceived need, covering all different stages from requirements gathering to validation of an experimental prototype.
As a result of literature review, the Dual Active Bridge topology was analytically explored and characterised. An additional degree of freedom was taken into account by the design of the high frequency transformer that will ensure galvanic isolation. A series of computational simulations are presented, in which the converter steady-state operation can be studied. A control loop intended for output voltage regulation is sized and tested for different load scenarios.
The experimental assembly of the DAB converter is described and, besides the outline of soft-ware and hardsoft-ware main features, emphasis is given to the implementation of two high frequency transformers based on previously presented designs. At last, practical trials are undertaken, and proof of concept is guarantee by an experimental prototype operating at 2 kW.
In the final analysis, proposals for future work are recommended so that additional efforts can lead to the accomplishment of even more promising results.
To begin with, I would like to thank Prof. Adriano Carvalho, who through his concise conversa-tions made me look to the engineering world in a completely different perspective.
A sincere gratitude towards everyone from AddVolt’s team shall be made, for taking me into their family. Special emphasis has to be given to Eng. José Pedro Araújo, for all the support, efforts, guidance and above all patience spent on a daily-basis during this months.
To the diurnal colleagues and to the nocturnal friends, notably Cordeiro, Schneider, Bandeira, Capa, Espassandim, Delfim, Maria João and all that remains to mention, I am truly thankful for making these five-years at FEUP the most pleasant experience possible. To the ones that paved my way towards this engineering degree, Zeza, Mota, Mafalda and Pacheco, I express my deeply gratitude.
For all the companion, understanding, constant support and recurring smile at the end of the day, I am absolute thankful to Betumes.
At last, for being my unconditional sponsors and the undoubtedly reason for this accomplish-ment, I owed you these last 23 years, and for that I leave the utmost thank you to my parents and sister.
Many thanks!
Nuno Alves
Oren Harari
Resumo i Abstract iii Acknowledgement v Abbreviations xvii 1 Introduction 1 1.1 Context . . . 1 1.2 Motivation . . . 2 1.3 Goals . . . 3 1.4 Document structure . . . 4
2 State of the Art 5 2.1 Bidirectional DC-DC Converters . . . 5
2.1.1 Non-isolated Bidirectional DC-DC Converters . . . 5
2.1.2 Isolated Bidirectional DC-DC Converters . . . 8
2.1.3 Equalizing topologies . . . 12
2.1.4 Market Research . . . 13
2.2 Semiconductor devices . . . 15
2.2.1 IGBTs . . . 15
2.2.2 Freewheeling diodes . . . 15
2.2.3 Wide Bandgap Semiconductors . . . 15
2.3 Modulation . . . 20
2.3.1 Pulse Width Modulation . . . 20
2.3.2 Phase shift voltage control . . . 21
2.3.3 Soft-switching . . . 21
2.4 Magnetics . . . 23
2.4.1 Basic principles . . . 23
2.4.2 Hysteresis loop and Core Materials . . . 24
2.4.3 Power losses . . . 26 2.4.4 Transformer modelling . . . 27 2.4.5 Design procedure . . . 31 2.5 Control . . . 35 2.5.1 Digital control . . . 37 2.5.2 Control technologies . . . 38 2.6 Batteries . . . 40 2.6.1 Types . . . 40 ix
2.6.2 Models . . . 42
2.6.3 Charging infrastructure . . . 45
2.7 Summary . . . 47
3 Dual Active Bridge (DAB) Converter Design 49 3.1 Characterisation and Steady-state Analysis . . . 50
3.2 Phase Shift Modulation . . . 52
3.3 Alternative Modulation Methods . . . 54
3.3.1 Extended Phase Shift . . . 54
3.3.2 Dual Phase Shift . . . 55
3.3.3 Triple Phase Shift . . . 56
3.4 Converter modelling . . . 56
3.5 Transformer design . . . 61
3.6 Summary . . . 68
4 Computational Analysis 69 4.1 Overview . . . 69
4.2 Open loop analysis . . . 72
4.3 Closed loop control . . . 76
4.3.1 The repercussion of digital delays . . . 80
4.4 System’s Response . . . 82
4.5 Summary . . . 83
5 Experimental Setup 87 5.1 Software development . . . 87
5.2 Hardware arrangement . . . 91
5.2.1 High frequency transformer . . . 92
5.2.2 Improved transformer design . . . 97
5.3 Summary . . . 102
6 Experimental Results and Discussion 103 6.1 Preliminary tests - proof of concept . . . 103
6.2 First improvements - modifications’ assessment . . . 105
6.3 System’s final validation . . . 106
6.4 Discussion . . . 109
7 Conclusion 111 7.1 Future Work . . . 111
7.1.1 Switching frequency . . . 111
7.1.2 Magnetics components . . . 112
7.1.3 Modulation and control . . . 112
A Core Datasheet 113
B AWG Chart 115
1.1 Overall system’s architecture . . . 3
2.1 Bidirectional Buck-Boost Converter . . . 6
2.2 Bidirectional Cascade Buck-Boost Converter . . . 6
2.3 Bidirectional Cuk converter . . . 7
2.4 Bidirectional Half-Bridge DC DC converter . . . 7
2.5 Typical structure of an Isolated BDC . . . 8
2.6 IBDC based on a voltage-fed full bridge and a current-fed full bridge . . . 9
2.7 IBDC based on a voltage-fed full bridge and a current-fed full bridge with active clamping . . . 9
2.8 Operating waveforms of an IBDC based on a voltage-fed full bridge and a current-fed full bridge with active clamping . . . 10
2.9 Dual Active Bridge . . . 10
2.10 Operating Waveforms of a Dual Active Bridge . . . 11
2.11 IBDC based on a voltage-fed half bridge and a voltage-fed full bridge . . . 11
2.12 Dual Active Bridge Series Resonant Converter . . . 12
2.13 Isolated DC-DC converter equalizing methods . . . 13
2.14 Technology Readiness Level . . . 14
2.15 Semiconductor energy band . . . 17
2.16 Sinusoidal pulse width modulation . . . 20
2.17 Phase shift voltage control waveforms . . . 21
2.18 Hard-switching losses . . . 22
2.19 Soft-switching current and voltage waveform . . . 22
2.20 Analogy between magnetic and electrical quantities . . . 24
2.21 B-H characteristic curve . . . 25
2.22 Two-winding transformer . . . 27
2.23 Transformer model with finite magnetising inductance . . . 29
2.24 Leakage flux in a transformer . . . 30
2.25 Total loss vs flux density . . . 32
2.26 Elementary feedback loop . . . 35
2.27 Electrochemical cell cross-section . . . 40
2.28 Rint Model equivalent circuit . . . 43
2.29 RC Model equivalent circuit . . . 43
2.30 First order model equivalent circuit . . . 44
2.31 Second order model equivalent circuit . . . 45
3.1 Dual Active Bridge topology . . . 50
3.2 DAB Lossless model . . . 51
3.3 Transformer voltages and inductor current for the phase shift modulation . . . 53
3.4 Alternative modulation: (a) Single Phase Shift, (b) Extended Phase Shift . . . 55
3.5 Alternative modulation: Dual Phase Shift . . . 56
3.6 Alternative modulation: Triple Phase Shift . . . 57
3.7 Simplified DAB converter structure . . . 57
4.1 Transferable power regarding different input voltages and phase shifts . . . 71
4.2 DAB converter’s power stage implemented in PSIM . . . 72
4.3 Phase shift modulation with dead time . . . 73
4.4 Phase shifted gate pulses with dead time . . . 74
4.5 Voltage at each transformer’s terminals and voltage and current at the leakage inductance . . . 74
4.6 Relationship between transferable power and phase shift . . . 75
4.7 Closed loop control diagram of the converter . . . 76
4.8 Plant’s Bode plot . . . 77
4.9 Plant’s unit-step response . . . 78
4.10 Plant’s root locus . . . 78
4.11 Bode plot of different DAB transfer functions . . . 78
4.12 Compensated system’s unit-step response . . . 79
4.13 Full order plant with digital delay root locus . . . 81
4.14 Compensated step response tuned by Ziegler-Nichols Second Method . . . 81
4.15 PSIM’s closed loop control block diagram . . . 83
4.16 Output voltage’s transient response for 15 kW to 0 kW load change . . . 84
4.17 Transferred power and output voltage’s transient response regarding a +15, -15, 0 kW load change . . . 85
4.18 Current flow for the +15, 15, 0 kW load variation test . . . 85
4.19 Discrete implementation of the generalised integrator . . . 85
4.20 Output voltage transient response with the PI+PR controller . . . 86
5.1 Basic principle of phase shifted PWM generation . . . 88
5.2 IGBTs in complementary switching . . . 89
5.3 IGBTs switching with dead time . . . 89
5.4 IGBT commutation delay . . . 89
5.5 Comparison between PWM digital signal and IGBT commutation . . . 90
5.6 State machine and flowchart of the developed code . . . 90
5.7 DAB prototype . . . 91
5.8 Implemented high frequency transformer . . . 93
5.9 Open circuit test - secondary winding kept open . . . 94
5.10 Open circuit test - primary winding kept open . . . 95
5.11 Short circuit test - secondary winding short circuited . . . 96
5.12 Short circuit test - primary winding short circuited . . . 97
5.13 Nanocrystalline core specifications . . . 97
5.14 Core saturation curve . . . 98
5.15 Improved high frequency transformer . . . 99
5.16 Transformer testing for parameters estimation . . . 100
5.17 Nanocrystalline core loss . . . 101
6.1 Experimental waveforms at 15 kHz . . . 104
6.3 Experimental waveforms at 30 kHz . . . 105
6.4 Input and output voltages and primary winding current at 2 kW . . . 106
6.5 Input and output measurements . . . 107
6.6 Voltages and current at the transformer . . . 108
6.7 System’s transient behaviour . . . 108
2.1 Comparison of different IBDC topologies . . . 14
2.2 Comparison between Si and WBG semiconductors material properties . . . 16
2.3 Breakdown voltages of p-n junction diodes based on different semiconductors ma-terials . . . 18
2.4 Drift region thickness of p-n junction diodes based on different semiconductors materials . . . 19
2.5 Terminology related to transformer design procedure . . . 33
2.6 Effects of increasing each parameter . . . 36
2.7 Comparison of different battery types’ specifications . . . 42
2.8 Charging modes defined in IEC 61851-1 . . . 45
3.1 High voltage full bridge possible states . . . 51
4.1 Assumed parameters for leakage inductance’s calculation . . . 70
4.2 Ziegler-Nichols Second Method Tuning Rules . . . 82
5.1 Real and estimated parameters of the developed high frequency transformer . . . 96
5.2 Improved transformer characterisation . . . 102
6.1 Outline of tests made . . . 103
Ac Cross-sectional Area
AC Alternating Current
Ah Ampere-hour
At Surface area
AWG American Wire Gauge B Magnetic Flux Density Br Residual Flux Density
Bs Saturation Flux Density
BDC Bidirectional DC-DC Converter BMS Battery Management System CCS Combined Charging System C Capacitor or Capacitance CPU Central Processing Unit
DC Direct Current
DOD Depth of Discharge DAB Dual Active Bridge DPS Dual Phase Shift
DSP Digital Signal Processors EMI Electromagnetic Interference
eV Electron Volt
EV Electric Vehicle
EVSQ Electric Vehicle Supply Equipment EPS Extended Phase Shift
F Magnetomotive Force
FPGA Field-Programmable Logic Array GaN Gallium Nitride
H Magnetic Field Intensity Hc Coercive Force
HEV Hybrid Electric Vehicle
Hz Hertz
i Current
iGSE Improved General Steinmetz Equation iLM Magnetising current
IBDC Isolated-Bidirectional DC-DC Converter IGBT Insulated Gate Bipolar Transistor
J Current density
k Coupling coefficient or SI prefix kilo Ku Window utilisation factor
L Inductor or Inductance Lm Magnetising Inductance
M Mutual Inductance
MLT Mean Length per Turn MnZn Manganese-Zinc
MOSFET Metal Oxide Semiconductor Field Effect Transistor
n Turn ratio N Number of turns NiCd Nickel-Cadmium Ni-MH Nickel-Metal-Hydride NiZn Nickel-Zinc Oe Oersted
PCB Printed Circuit Board PFC Power Factor Correction
PID Proportional–Integral–Derivative PR Proportional-Resonant
PWM Pulse Width Modulation R Resistance or Resistor
R Reluctance
RAM Random-Access Memory
RCD Resistor-Capacitor-Diode
ROM Read Only Memory
Si Silicon
SiC Silicon Carbide SOC State of Charge SOH State of Health SPS Single Phase Shift
T Tesla
TPS Triple Phase Shift
TRL Technology Readiness Level VSI Voltage Source Inverter
WBG Wide Bandgap
ZCS Zero Current Switching ZVS Zero Voltage Switching
δ Skin depth
φ Magnetic flux
ϕ Phase shift or Phase-delay µ Permeability or SI prefix micro
Introduction
This dissertation will provide a thorough study of bidirectional isolated DC-DC converters, cov-ering all the stages from specifications gathcov-ering to prototype testing and validation. This first chapter introduces the subject under study, providing a framework for the increasing need of such a power flow.
The motivation that has led to the commission of this thesis is presented, along side with the respective objectives that are expected to be fulfilled. Before diving into the main report, a high-level review of the overall document is summarised.
1.1
Context
Nowadays, environmental sustainability is one of the most critical global concerns. Unarguably, greenhouse emissions generated by the transportation sector are one of the major sources of pol-lution released into the atmosphere. As a result, automotive industry is facing a paradigm shift with the sharp rising of hybrid and full electric vehicles. Factors such as consumer acceptance, governments incentives, novel technologies and increasing environmental awareness are the main drivers of this revolution.
In parallel, power electronics’ role does not only concern vehicle traction but is also related to various new power conversion applications. A fundamental part widely observed in electric vehicles is a DC-DC converter. This electronic block is used for highly diverse purposes, ranging from interfacing battery banks with the electric powertrain to supplying energy to auxiliary loads. In traditional DC-DC converters, power flow occurs only in one direction. However, adjustments can be made in order to enable power transfer in both ways [1].
Recent developments in bidirectional DC-DC converters have been triggered by the need of energy transfer between two DC buses within a system. Besides, there is a wide variety of areas which share this common need of bidirectional power flow. The evidence of bidirectional power transfer can be clearly seen in the case of a photovoltaic system; these applications are typically connected to a storage unit for constant load powering, thus a bidirectional DC-DC converter is required for both charging and discharging the battery bank - during charging process, energy is
supplied to the battery from the photovoltaic panel. When there is a need to power the loads with the stored energy in the battery bank, power flows occurs in the reverse way through the DC-DC converter. Electric and hybrid vehicles demands a DC-DC converter with bidirectional power flow capability to be interfaced between the battery bank and the electric motor. Uninterruptible power supplies also require a bidirectional converter to enable their two distinct modes of operation.
Even though non-isolated bidirectional DC-DC converters exist, galvanic isolation is usually a requirement, supported by the need of noise reduction, safety and standards-compliance. A high frequency transformer is used to achieved these prerequisites, while at the same time enabling the voltage matching expected in numerous applications.
1.2
Motivation
Having stated the importance of new trends in DC-DC converters in today’s power electronics market, it should now be presented why there is an urge to develop such a converter.
Given the fact that this dissertation is being developed in a company environment, a brief introduction of the enterprise is to be made. This thesis has been commissioned by AddVolt - this startup founded by FEUP’s alumni has developed a technology that allows the recovery, production and the storage of electrical energy on-board of trucks. This energy can therefore be used in the refrigeration process of the transported goods as an alternative to diesel fuelled systems [2].
Generally, a refrigerator truck demands two diesel engines, the first for vehicle’s traction and the latter for the cooling process. AddVolt’s solution combines two different energy sources; be-sides from photovoltaic energy generated by placing photovoltaic panels on the truck’s ceiling, energy is also seized from braking - regenerative braking - therefore allowing to power the re-frigerator in electrical mode. In consequence, this on-board energy production enables haulage companies’ operating costs reduction and decreases overall fossil fuels consumption impact. In addition, the generated energy can be shared with other trucks, as well as it can be injected in the utility grid.
Preliminary results show that an 87% reduction in carbon dioxide emissions can be achieved, along with a 30dB noise reduction, representing a total of 30%. Fuel consumption together with vehicle’s maintenance needs is sharply diminished.
Above all, the requested bidirectional DC-DC converter is, therefore, to be interfaced between the battery-bank and the 700V DC bus of the DC-AC inverter used either to power the motor or to connection to the power grid. Figure 1.1 represents the overall system’s architecture where the proposed isolated bidirectional DC-DC converter (IBDC) is to be connected. This new converter emerges as a replacement for a traditional buck-boost being used. Galvanic isolation is a necessity imposed mainly for safety purposes, standards-compliance and costumer’s demands.
Figure 1.1: Overall system’s architecture
1.3
Goals
The fundamental ambition of the proposed work is to develop an isolated bidirectional DC-DC converter. Given the intricacy of the subject under study, the concept of the work to be accom-plished must be well-defined. In addition, a proper planning of the different work’s phases ought to be made.
With this dissertation, a 15 kW isolated bidirectional DC-DC converter is expected to be im-plemented. In broad terms, the fundamental milestones guiding the work can be listed as:
• Literature review should be made on pertinent topics that provide added value to the con-verter desired to be implemented. Based on the requirements established, a suitable topol-ogy is to be chosen.
• Further study on the selected topology ought to be followed by converter’s design, with emphasis on the high frequency transformer sizing and characterisation.
• Simulations of the converter and complementary control loop should be undertaken to facil-itate the subsequent experimental realisation.
• An experimental setup should be arranged and validation of basic software and hardware features is to be reported.
• Verification of the developed prototype must be investigated through several tests under substantial power levels.
In conclusion, the final dissertation should be a culmination of a study, development, imple-mentation and validation of an isolated bidirectional DC-DC converter which shall comply with the requirements set out by those involved. On a personal level, it is expected that a wide range of skills regarding system’s engineering, in particular related to the field of power electronics, are attained and mastered throughout this opportunity along side with a company focused on that area.
1.4
Document structure
This document is organised as follows:Chapter 1 introduces this dissertation, by providing the context in which its exist, as well as the motivation behind this work. Thesis’ main goals are listed, and an outline of the document structure is also provided.
Chapter 2 provides a literature review in the area of isolated bidirectional DC-DC converters, regarding their topology and modulation techniques. A brief analysis of semiconductors power devices is also carried out, since there is an interest in new technologies that aim to achieve better system’s performance. Since a major requirement is galvanic isolation, a revision of elementary magnetic principles allied with transformer modelling and design is performed. Digital control and different control techniques are also covered. For better understanding of converter’s low voltage DC bus, several concepts in the area of batteries are presented.
Chapter 3 presents a detailed characterisation of DAB converter principle of operation. Dif-ferent modulations strategies are studied along with the design of the high frequency transformer. Chapter 4 explores converter’s behaviour by computational simulations. A control loop in-tended to regulate output voltage is developed and tuned. Analysis of system’s response to differ-ent load scenarios is given.
Chapter 5 unveils the implemented prototype by characterising its inherent software and hard-ware. Focus on the high frequency transformer is given and its parameters are estimated.
Chapter 6 covers the obtained results from the experimental setup assembled during all dif-ferent stages of testing. A critical assessment of the outcome retrieved is performed.
Chapter 7 closures the dissertation and suggests directions for future work in bidirectional isolated DC-DC converters.
State of the Art
Nowadays, there has been an increasing debate and concern regarding the sustainability of the planet. Efforts from both society and scientific community have been urging governments to take action in order to preserve the environment and, as it could be verified in 2015 at the Paris climate conference, the world has come together on a joint fight for greenhouse gases emissions mitigation. At the same time, power electronics play a key role in this movement, being a crucial element in renewable energy systems and in electric vehicles (EV). Therefore, this section will analyse bidirectional DC-DC converters, dividing them according to the eventual presence of isolation. Since high performance is required, semiconductor devices and soft-switching techniques will also be studied. Since galvanic isolation calls for a high-frequency transformer, a revision of magnetic principles will be considered. A brief analysis of concepts regarding digital control and batteries will also be presented.
2.1
Bidirectional DC-DC Converters
Researches have been focusing on Bidirectional DC-DC converters (BDC) due to the growing need of energy transfer between two DC buses within a system; renewable energy generation systems are usually connected to a energy storage device in order to continuously power loads, hence a DC-DC converter with bidirectional power flow capability is required. On the other hand, electric vehicles employ BDC to establish the connection between batteries and the electric motor. BDC can be categorised into non-isolated and isolated converters and will be further analysed in the following sections.
2.1.1 Non-isolated Bidirectional DC-DC Converters
Essentially, a non-isolated BDC can be obtained from the common unidirectional DC-DC con-verters, by improving the unidirectional power flow capability of the conventional converters with bidirectional switches. Therefore, non-isolated bidirectional DC-DC converters usually share a simple structure, both high efficiency as well as reliability, among other advantages.
Considering a classic boost converter, the presence of a diode hinders a bidirectional power flow. However, by replacing the diode with a transistor - with an eventual anti-parallel diode -, current conduction in both directions becomes possible, thus allowing bidirectional power flow.
Various non-isolated BDC have been reported in literature [3]. The simplest topology can be obtain by swapping the diode in a typical buck boost topology with a bidirectional switch as seen figure 2.1.
Figure 2.1: Bidirectional Buck-Boost Converter - adapted from [4]
Both switches are supposed to never conduct at the same time, therefore a small dead time should be implemented. In step-up operation, transistor Q1 conducts accordingly to a specific
duty cycle, whereas Q2is kept open. On the other hand, during step-down operation, switch Q1is
kept off while Q2conducts.
An alternative topology can be derived by cascading both the bidirectional buck and boost con-verter, as seen in figure 2.2. Depending on transistors’ commutation combinations and direction of the current flow, output voltage can be higher or lower than the input voltage [4].
Figure 2.2: Bidirectional Cascade Buck-Boost Converter [3]
For the step-down operation in forward direction, only switch Q1needs to be operated; while
the load. When switch Q1commutates to the off state, D4is direct biased, and inductor’s storage
energy powers the load.
Still in forward direction, in step-up operation, transistor Q1is always conducting whereas Q2
commutates with variable duty cycle - usually in motoring applications depends on motor speed. Both switches Q3 and Q4 are off and diode D4 is reversed biased. Inductor is supplied by the
battery while the load is powered by capacitor C2when switch Q2in on. When Q2turns off, diode
D3is direct biased then both load and capacitor C2receive energy from the inductor.
Reverse power flow can occur in different scenarios depending on voltage levels of the battery and load. If load’s voltage is higher than the battery voltage, the converter operates in step-down mode. Therefore, Q3 commutates at a fixed switching frequency with variable duty cycle, while
Q1, Q2 and Q4 are kept off. During an entire switching period, D4is reversed biased and D1 is
direct biased. As load’s voltage level decrease below battery voltage, the converter operates in step-up mode, thus Q3is always conducting, whereas transistor Q4commutates with variable duty
cycle. In this mode of operation, both Q1and Q2are always off.
Another topology for bidirectional power flow can be obtained by replacing the diode on a conventional Cuk converter by a bidirectional switch [3], as seen on figure 2.3.
Figure 2.3: Bidirectional Cuk converter [3]
At last, figure shows a non-isolated bidirectional half-bridge DC-DC converter. Accordingly to switches commutation configuration, both buck and boost operation can be obtained.
Figure 2.4: Bidirectional Half-Bridge DC DC converter - adapted from [3]
When compared to Cuk converter, this topology has the advantage that only one inductor is needed, hence reducing the number of passive elements. Efficiency is also usually higher since inductor is subject to lower current values, hence lower conduction and switching losses [3].
2.1.2 Isolated Bidirectional DC-DC Converters
Noise reduction, protection and personnel safety are the main factors why several standards im-pose galvanic isolation in bidirectional DC-DC converters [1]. In contrast, voltage matching is also required in various applications; thus a magnetic transformer can be used to achieve these requirements.
A typical Isolated-Bidirectional DC-DC Converter (IBDC) can be seen in figure 2.5. This structure consists of a high frequency inverter, a high frequency transformer and a high frequency rectifier. This transformer is used for both galvanic isolation and voltage matching; since it requires alternating current at its terminals, an inverter is used on each side. Moreover, this structure requires that DC buses must be able to either generate or absorb energy, which results in connecting them to an active load such as a battery [1].
Figure 2.5: Typical structure of an Isolated BDC [1]
Even though all IBDC share the previous structure presented, it is possible to classify topolo-gies as current-fed, voltage-fed and as a combination of them [5]. A voltage-fed structure has a capacitor at its terminals, like a common buck converter, acting as a voltage source. A current-fed, or current type, IBDC structure has an inductor at its terminals, acting as a current source. In a voltage-fed manner, the converter can be directly connected to the DC source or battery bank, whereas in a current-fed structure, an additional DC inductor with stiff current characteristics must be employed in between.
One of the broadly used topologies is the current-fed and voltage-fed full bridges. In this structure, figure 2.6, one full bridge is connected directly to the DC source - voltage-fed - and the other is a current-fed full bridge, connected to the DC inductor.
Nonetheless, this topology is not reasonable for medium-high power applications because the leakage energy stored in the transformer causes high voltage spikes on switches during switching in the current-fed side since it has to find a path to discharge. Different derivations have been develop, such as the introduction of active clamping, RDC and lossless snubbers enabling the usage of this topology in high power applications [5]. Figure 2.7 shows a derivation where the addition of an extra switch and capacitor to the current-fed side acts as active clamping [6].
Figure 2.6: IBDC based on a voltage-fed full bridge and a current-fed full bridge [5]
Figure 2.7: IBDC based on a voltage-fed full bridge and a current-fed full bridge with active clamping [6]
Bearing in mind the terminology presented in figure 2.5, in mode A to B, the bridge diagonal switching pairs operate at duty cycles larger than 50%. During the overlapping period, when all four bridge transistors are conducting, the input inductor is charged. When one diagonal switch pair is turned off, the inductor begins to discharge and the clamp transistor begins to conduct. Mode A to B operating waveforms can be seen in figure 2.8.
In parallel, when the voltage-fed side is the active converter, a phase-shift modulation is used, enabling zero voltage switching (ZVS) for all transistors.
This current-fed topology is suitable for battery operated applications and power factor cor-rection (PFC), since is has a relatively low ripple input current. By contrast, a few disadvantages can be listed such as bulky input side inductor and the previously mentioned voltage spikes due to the transformer leakage inductance; even though this impact can be attenuated through active or passive clamping, a higher element count arise as a consequence.
By contrast, another topology widely used for bidirectional converters is the Dual Active Bridge (DAB), as shown in figure 2.9. In this topology, two voltage-fed full bridges are directly connected to the DC sources and the control is based on phase-shift modulation; a nearly square wave AC voltage is supplied across the transformer terminals by turning on the diagonal switching pairs simultaneously with a 50% duty cycle and with 180 degrees phase shift between two legs.
Figure 2.8: Operating waveforms of an IBDC based on a voltage-fed full bridge and a current-fed full bridge with active clamping [6]
This phase shift determines the direction and amount of power flow between the two DC buses -its control enables a fixed frequency operation with full control over the transferable power.
Figure 2.9: Dual Active Bridge [5]
Figure 2.10 show the main operating waveforms of this topology. Considering the terminology presented in figure 2.5, if vac,Aleads vac,B, the phase-shift ϕ is considered positive and thus the
power flow is from side A to side B. On the other hand, to transfer power from side B to A, vac,A
should lag vac,B, hence ϕ is negative. This phase-shift is implemented by proper timing control of
each converter switches.
Various advantages can be listed for this dual voltage-fed topology such as high efficiency, high power density, zero voltage switching achievable in both bridges associated to flexible and simply control strategies. However, it has been referenced that under light load conditions high
Figure 2.10: Operating Waveforms of a Dual Active Bridge [1]
efficiency becomes hard to achieve [7]. Narrow voltage range for optimal operation emerges as other disadvantage of this topology.
Several derivations of DAB have been developed, such as the combination of two voltage-fed half bridges. This structure only uses half the number of devices as the full bridge converter, however, they are subject to twice the DC input voltage; while in some applications this may seem an advantage because of the voltage ratings of the low voltage side, in medium-high power applications it becomes a huge disadvantage. These derivations are flexible, since they allow the combination of different structures on each side of the high-frequency transformer, as shown in figure 2.11; in this topology, DAB converter was modified to a combination of a voltage-fed half bridge and a voltage-fed full bridge.
Figure 2.11: IBDC based on a voltage-fed half bridge and a voltage-fed full bridge [5]
Moreover, a series-resonant circuit can be implemented, allowing natural zero current switch-ing (ZCS) by simply adjustswitch-ing the transformer operatswitch-ing frequency to the resonant frequency. Figure 2.12 shows a DAB converter where the two full bridges are interconnected through a series
LC resonant tank and the high-frequency transformer.
Figure 2.12: Dual Active Bridge Series Resonant Converter [8]
In this topology, setting the switching frequency higher than the LC resonant tank allows the converter to operate only in continuous current mode. The LC tank also works as a bandpass filter, thus blocking DC current components, preventing the transformer from saturation. Furthermore, ZVS and ZCS allows the increasing of switching frequency, hence magnetic components can be reduced [8].
2.1.3 Equalizing topologies
As seen in figure 1.1, a battery pack is usually utilised for propulsion in EV and HEV applica-tions. These packs consist of series-connected batteries for high voltage achievement, which are subjected to imbalance problems. As a result, a significant reduction in batteries’ lifespan and potentially permanent damage in the constituent cells may occur. In this way, several equalizing methods can be found in the literature [9]; however due to the focus of this thesis, only the ones regarding bidirectional power flow combined with galvanic isolation will be reviewed.
Accordingly to [9], non-dissipative methods based on IBDC can be categorised into two types. In a distributed isolated DC-DC converter structure, as the one seen in figure 2.13a, every cell in a series-connected battery pack has a dedicated isolated flyback converter. Bidirectional power flow between each cell and the battery pack can be verified, therefore an advantage that the unidirec-tional derivation does not possess.
On the other hand, in these topologies, active and magnetic element count is directly pro-portional to the number of cells connected in series. Reference [10] presents an experimental setup scaled for four series-connected battery modules, thus resulting in a solution with a total 32 switches.
For the purpose of reducing element count, an equalizing method based on a centralized iso-lated DC-DC converter can be implemented. As seen in figure 2.13b, a number of identical sec-ondary coils, equal to the number of cells, are connected to a common core. As a result, the number of active elements is reduced, since energy can be directly transferred from the most charged cells to the undercharged ones through the shared transformer. A drawback of this topology is the diffi-culty in achieving a transformer with many secondary coils for a single core for applications with large battery packs [9].
(a) Distributed isolated DC-DC converter structure
(b) Centralized isolated DC-DC converter structure
Figure 2.13: Isolated DC-DC converter equalizing methods [9]
Having study different IBDC topologies, a advantages vs disadvantages analysis should be made in order to summarise all the information retrieved - table 2.1 lists the referred comparison. It should be noticed that full-bridge converter with series resonant tank and charge equalizing topologies where not taken into account in this analysis since their inherent high element count, resulting in non-compact solutions, and complex control strategies.
2.1.4 Market Research
Being developed in an entrepreneurial environment, it is of utmost importance that a market anal-ysis on bidirectional isolated DC-DC converters is made. By this means, it is possible to identify costumer’s needs, to find potential competitors and to determine how the company’s product will distinguish from another options, namely efficiency, cost, size and performance.
With that in mind, an attempt to identify main manufacturers and their key products was under-taken. Firstly, the France-based power electronics dedicated Tame-Power offers general purpose DC-DC converters within the kW range - up to 60 kW -, with the capability of bidirectional power flow [11]. Even though galvanic isolation is not available off-the-shelf, such feature can be in-cluded upon request.
Although at a slightly lower power rating, Creative Power Technologies possesses a 5 kW bidi-rectional isolated battery charger [12]. Main characteristics can be listed as up to 93% efficiency, protection level of IP20 and communication over a Modbus interface.
Through a press release, TDK-Lambda has announced the development of a 11 kW modular bidirectional isolated DC-DC converter [13]. Claiming efficiency up to 95%, this converter can be paralleled 5 times, hence reaching a power level of 50 kW. It is worth mentioning the high power
Table 2.1: Comparison of different IBDC topologies - adapted from [1]
Topology Advantages Disadvantages
CF-FB
- Inherent protection against over current and short circuit - Relatively low ripple input current, suitable for PFC or battery operated applications
- Difficult start-up procedure, requires extra circuitry - Voltage spikes due to transformer leakage inductance, implies high losses in high-freq applications - Bulky input side inductor - High-ripple output current, implies high quality capacitor - High transient voltage at LV side, implies active or passive clamping, higher element count
DAB (VF-FB)
- Voltage stress limited to DC bus level - No need for additional elements for soft-switching
- Very flexible control - ZVS can be achieved in both bridges - Number of switches lower than CF-FB
- High efficiency
- Lost of soft-switching in light load conditions - High number of switches, results in larger driver size, higher gate losses and increase overall cost - High circulating current flow when voltage ratio isn’t 1
CF-DHB
- Intended for fuel-cell and battery applications - Minimum count of switches - ZVS achived without additional elements
- Reasonable ripple DC current at LV side
- Switching devices are subjected to twice DC voltage of the battery, so may not be suitable for connection with higher battery voltages -Split DC capacitors have to withstand the rated current value - Voltage balancing circuits may be required - Unbalanced current stress between two switches in the LV side - Lack of reference to medium-high power applications
density achieved by the Japanese company, whose converter slim dimensions make it suitable to be mounted on a 1U rack unit.
At last, Bel Power Solutions, Inc holds this dissertation’s main competitor, the 700BDC150 series DC-DC converter [14]. This converter is suitable for up to 15 kW power, guaranteeing full galvanic isolation between each ports. An interface with CAN Bus can be found allied with a stated efficiency close to 97%. Besides that, an IP67 protection degree is presented and cooling can be achieved by either liquid or convection. By far, this is the product that matches the most the requirements and characteristics of the converter expected to be implemented in this thesis.
An interesting way of comparing the outcome of this dissertation with the abovementioned products is by the Technology Readiness Level (TRL) method. This measurement scale proposed by NASA is used to estimate the maturity of a given technology and can be summarised as follows in figure 2.141. Although competitors’ systems are already on the last level, this dissertation should strive to ensure a high position on the TRL ladder.
Figure 2.14: Technology Readiness Level
2.2
Semiconductor devices
EV and Hybrid electric vehicles (HEVs) applications strongly rely on power semiconductors de-vices. Components such as insulated gate bipolar transistors (IGBTs) and freewheeling diodes represent a key role in various power electronics systems, ranging from traction’s inverters to on-board DC-DC converters. Therefore, semiconductors devices impose overall system’s cost, performance and efficiency. In consequence, HEV and EV market growth serve as a driving force for emerging semiconductors technologies.
2.2.1 IGBTs
Nearly all power electronics converters in a EV and HEV employ IGBTs as the main switching element, mainly due to their superior current conduction capability in comparison to Power Metal Oxide Semiconductor Field Effect Transistor (MOSFETs) [15].
Besides sharing similar operation and structure with MOSFETs, IGBTs assure a reduction in on-state conduction losses through conductivity modulation; even though a likewise high input impedance and fast turn-off speed exists, a better current carrying ability and on-state voltage drop are verifiable. For this reason, when conduction losses minimisation must be granted in high-voltage systems, IGBTs appear as a better alternative to traditional Power MOSFETs.
2.2.2 Freewheeling diodes
Even though being used mostly as anti-parallel freewheeling devices, diodes play a key role in power converters. Having a direct influence in system’s performance, voltage and current ratings, forward voltage drop and recovery characteristics are the main parameters that should be taken into account in selecting diodes. Both turn-on and turn-off transitions, reverse and forward bias respectively, demand a limited amount of time to occur.
In hard switching conditions (see section 2.3.3), IGBT turn-on losses are proportionally related to the increase of reverse recovery process and long recovery time. Therefore, fast and soft recov-ery characteristics are features that freewheeling diodes should have in order to reduce switching losses; both a diminished peak recovery current and didt are qualities that freewheeling diodes for high performance applications should have.
2.2.3 Wide Bandgap Semiconductors
On-board electronics systems are subjected to severe operating conditions. Under those circum-stances, temperature arises as the most adverse condition for power electronics, as a result of the produced heat by the engine, the environment and electrical devices losses. Presently, semicon-ductors devices are based on Si materials, whose maximum junction temperature limit is around 150oC. For this reason, power electronics must be combined with a cooling mechanism, in order to maintain power devices’ temperature below the mentioned limit.
At the same time, a heatsink must be employed for electronics’ cooling process. Usually, a heatsink fills up to a third of the total converter volume, therefore increasing overall system’s cost, size, weight and volume [16]. As a result, the development of power devices with higher temperatures ratings would diminished the need of a complex cooling mechanism, as well as a consequent overall system size and cost reduction. Nevertheless, Si devices have reached their maximum temperature limits, thus an attempt to improve their performance is not feasible since Si technology is being operated at its blocking voltage, temperature and switching frequency limits [17].
Provided that, recent developments have been focusing on Wide Bandgap (WBG) semicon-ductors devices to overcome Si’s physical limitations. Among the various candidates, Silicon Carbide (SiC) and Gallium Nitride (GaN) present better features such as higher blocking voltage capability, higher operating temperature and higher switching frequencies than traditional Si ma-terials. Table 2.2, adapted from [16], summarises a few properties of WBG materials; some of this properties will be explained next.
Table 2.2: Comparison between Si and WBG semiconductors material properties - adapted from [16]
Property Si 6H-SiC 4H-SiC GaN Diamond Bandgap, Eg(eV) 1.12 3.03 3.26 3.45 5.45
Dielectric constant, εr 11.9 9.66 10.1 9 5.5
Electric breakdown field, Ec(kV/cm) 300 2500 2200 2000 10000
Electron mobility, µn(cm2/V·s) 1500 500 1000 1250 2200
Hole mobility, µp(cm2/V·s) 600 101 115 850 850
Thermal conductivity, λ (W/cm·K) 1.5 4.9 4.9 1.3 22 Saturated electron drift velocity, vsat(×107cm/s) 1 2 2 2.2 2.7
ε = εr· εo, where εo= 8.85 × 10−14F/cm
From a brief analysis of the previous table content it can be concluded that GaN devices are best suitable for high-frequency, low power applications, due to their high electron mobility and poor thermal conductivity. On the other hand, SiC is appropriated for high power applications as a result of high thermal conductivity. Even though GaN technologies show overall better perfor-mance characteristics than SiC devices, the latest are more likely going to be able to replace Si devices in future electronics market; this assumption is made under the fact that there is a known difficulty in finding good quality substrates required for these devices’ manufacture [17].
Based on table 2.2, a more detailed explanation of the most important properties will follow.
2.2.3.1 Bandgap
In solid-state physics, electrons exist at several energy levels that when are combined form an energy band. A semiconductor energy band structure can be verified in figure 2.15. A bandgap usually refers to the region between the top level - conduction band - and the lower level - valence band - where ideally no electrons can exist [16].
Figure 2.15: Semiconductor energy band
By being externally excited, electrons in the valence band can transit to the conduction band, therefore becoming a conduction electron able to behave as a electrical charge carrier. Therefore, bandgap can be defined as the energy difference, measured in electron Volt (eV), between both tops bands, thus resulting in bandgap Eg= Ec− Ev.
Since electrons required a specific amount of energy to move from the valence to the conduc-tion band, it is expected that this energy difference depends with distinct materials properties. In a conductor, this prohibited region does not exist, hence a overlap between energy bands occurs, thus enabling excellent electric current conductivity. On the other hand, in a isolator material, a large bandgap is present, so a high amount of energy is needed in order to allow electrons’ transi-tion from the valence band to the conductransi-tion band. A semiconductor is therefore a material with a more reduced bandgap than an isolator. Considering table 2.2, Si has a bandgap Egof 1.12 eV,
while WBG semiconductors as SiC and GaN present bandgap three times higher than Si.
In parallel, as the temperature increases, electrons at the valence band become more agitated. By exceeding a certain temperature, electrons can uncontrollably move from the lower level to the conduction band; as previously mentioned, this boundary temperature is around 150oC for Si materials. For WBG semiconductors, since their bandgap is longer, the energy required to allow electrons transitions is consecutively higher, hence higher temperatures can be withstood; therefore, WBG devices can be employed in applications that surpass Si’s physical limitations. Accordingly to [15], SiC and other WBG semiconductors can operate to temperatures as high as 500oC.
2.2.3.2 Electric Breakdown Field
Despite having discussed the benefits of a wider bandgap presence in a semiconductor, it is also important to notice that wider bandgap results in a higher electric breakdown field, which in power electronics translates in devices with inherent higher breakdown voltages.
Accordingly to [18], the breakdown voltage of a non-punch through - p-n junction - diode can be expressed as
VBD≈
ε · Ec2
2qNd
where ε is the absolute permittivity, Ec the electric breakdown field, q is the electric charge
and Ndthe doping density.
By using material properties from table 2.2, the breakdown voltage of silicon yields
VBDSi ≈11.9 · 8.85 × 10 −14· 300 × 103 2 · 1.602 × 10−19Nd (2.2) which results in VBDSi ≈2.96 × 10 17 Nd (2.3)
Applying equation 2.1 to WBG semiconductors materials from table 2.2 results in the follow-ing estimates
Table 2.3: Breakdown voltages of p-n junction diodes based on different semiconductors materials
Material Breakdown Voltage Normalisation to Si
Si 2.96×1017/Nd 1
4H-SiC 1.35×1019/Nd 46
6H-SiC 1.67×1019/Nd 56
GaN 9.94×1018/Nd 34
Diamond 1.52×1020/Nd 514
It should be noticed that table 2.3 was obtained by assuming a constant doping density for all materials. As it can be verified in the third column of the previous table, much higher breakdown voltages can be achieved by WBG semiconductors. In addition, more doping density can be injected in the semiconductors material due to the higher electric breakdown fields, thus increasing even more the breakdown voltage limit of this new emerging technology.
Additionally, a reduction in the drift region thickness can be obtained as a result of higher electric breakdown field and higher doping density. Reference [18] relates the depletion layer width with the corresponding breakdown voltage as
W(VBD) ≈
2VBD
Ec
(2.4)
By using the formerly obtained values of table 2.2 and 2.3, the drift region thickness of differ-ent semiconductors materials can be calculated
By analysing the last column of table 2.4, it can be verified that thinner depletion layer widths can be obtained in WBG semiconductors, as a result of their high electric breakdown field.
At last, accordingly to [16], as higher breakdown voltages are achieved, doping levels can be increased, therefore WBG semiconductors’ on-resistance of the drift region is around 10 times smaller than the Si counterparts; it should be noticed that the on-resistance is inversely proportional to the electric breakdown field.
Table 2.4: Drift region thickness of p-n junction diodes based on different semiconductors materi-als
Material Drift region thickness Normalisation to Si # times thinner Si 6.67×10−6·VSi BD 1 -4H-SiC 0.91×10−6·V4H−SiC BD 0.14 7 6H-SiC 0.80×10−6·V6H−SiC BD 0.12 8 GaN 1×10−6·VGaN BD 0.15 6 Diamond 0.2×10−6·VDiamond BD 0.03 33 2.2.3.3 Drift velocity
As WBG semiconductors drift velocity are at least twice as higher than the Si material (see ta-ble 2.2) and since switching frequency is directly proportional to the material’s drift velocity, it is therefore expected that WBG devices are capable of being commutated at higher switching frequencies.
Furthermore, lower reverse recovery current and time can be attained as a result of the faster removal of charge in the depletion layer achieved by higher drift velocities [16].
2.2.3.4 High thermal conductivity
As seen in table 2.2, WBG materials present a higher thermal conductivity than Si based power devices. As stated in [16], higher values of thermal conductivity imply a lower junction-to-case thermal resistance, thus the generated heat can be dissipated more easily. Recalling table 2.2, GaN thermal conductivity of 1.3 W /cm · K is its major limitation - parameter even lower than the one observed in Si -, hence it usage in high-power applications is compromised.
2.3
Modulation
Recalling figure 2.5, it can be verified that a typical isolated bidirectional DC-DC converter struc-ture relies on two DC-AC converters connected through a high frequency transformer. Literastruc-ture review presented in 2.1.2 shows that most IBDC topologies are based on H-bridge voltage source inverters (VSI); several topologies regarding half-bridge structures were identified, however, for simplicity, during this section, focus will be given only to H-bridge inverters.
Accordingly to [18], a static power converter like the H-bridge inverter generates constant output voltage levels. As a result, so that an arbitrary voltage waveform can be generated, some kind of control ought to be applied to the inverter; by alternating the available voltage levels in such a way that the fundamental component of the output voltage waveform approximates the desired reference. This method is the so called modulation, which can be carried out by different techniques.
2.3.1 Pulse Width Modulation
Pulse Width Modulation (PMW) is the basis of modulation schemes in power electronics. Even though different techniques exist, a common premise based on the change of width of a high frequency switching pulse train so that the low frequency component of the generated voltage equals the desired reference is shared among them.
A rather common modulation scheme is the sinusoidal PWM, frequently also known as carrier-based modulation. A brief description of its operation can be provided, in which the desired output voltage is generated by the comparison of the reference waveform - known as modulator -, with a high frequency triangular carrier signal. Gate signals are determined by whether the modula-tor is larger or smaller than the carrier, thus generating an output voltage with a low frequency component replica of the modulator desired waveform.
Figure 2.16: Sinusoidal pulse width modulation [18]
Figure 2.16 represents the mentioned switching waveform generation in sinusoidal PWM. When the sinusoidal modulator signal is over the high frequency triangular wave, the gate signal is set to logic "1", which translate in a switch being turned on. Main advantages of this derivation
on modulation rely on the ease of implementation allied with a good power quality. Nevertheless, it has been reported that a trade-off between power losses and filter design should be conducted, since high switching frequencies result in reduction of system’s efficiency due to proportional in-crease of switching losses with the switching frequency. On the other hand, lowering the switching frequency results in bigger and bulkier filters, thereby justifying the need of a compromise between these two factors [18].
2.3.2 Phase shift voltage control
In phase shift modulation, control of the output voltage waveform of the H-bridge inverter can be achieved by phase shifting each half-bridge legs; both legs operate under square-wave mode with a fixed 50% duty cycle, yet a phase delay ϕ is imposed between the two legs. As a result, H-bridge’s output voltage surges as the difference of these two waveforms.
To give an illustration, phase shift voltage control can be represented in figure 2.17, where S1 and S2 are gate signals for each half-bridge leg and Vo is the inverter’s output voltage. By
introducing a phase lag between the two legs, output voltage becomes the difference between the two square-wave voltages at each leg’s output.
Figure 2.17: Phase shift voltage control waveforms
In contrast to PWM, the output voltage of a H-bridge inverter is not characterised by its low frequency average value, thus energy is transferred at the switching frequency, thereby allowing the employment of smaller magnetic components [19].
2.3.3 Soft-switching
Active components’ losses play the main role in a converter’s efficiency degradation. Switches’ losses can be divide in conduction losses - consequence of the on-resistance of the switch - and switching losses. These losses result during turn-on and turn-off transitions, as a consequence of an overlap period between voltage and current when the switch is commutating.
Figure 2.18 illustrates hard-switching; when the switch starts conducting, current rises before voltage drops. On the other hand, when turning off, voltage rises before current drops hence, this overlap losses translates in efficiency’ reduction, therefore limiting the maximum frequency of operation and power density [20].
Figure 2.18: Hard-switching losses [20]
This overlap energy losses can be minimised by increasing the rate of change of current and voltage, didtanddvdt respectively. However, fast switching causes voltage spikes and electromagnetic interference (EMI) to be generated, so an optimisation of dtdi anddvdt should be taken into account.
Soft-switching emerges as a good technique for improving efficiency by reducing switching losses, by trying to prevent or minimise the overlap period. In sum, soft-switching starts be setting an electrical parameter to zero before the switch commutation. For ZVS operation, in order to reduce turn-on losses, the switch is turned in only when voltage falls to zero, therefore eliminating any overlap with current. For ZCS operation, same principle applies, however commutation only occurs when current, instead of voltage, reaches zero, thus reducing turn-off losses.
Figure 2.19: Soft-switching current and voltage waveform [20]
Since there is no assurance that the switch has dissipated all its energy before commutation, usually a fast body diode is placed in parallel with the switch to ensure all energy will be drained [20].
2.4
Magnetics
Since galvanic isolation is a requirement, a study of power transformers and elementary electro-magnetic principles ought to be made. For this reason, based on the bibliography in [21, 22], this section will cover some elementary principles of magnetism as well as a design procedure for transformers; it it expected that the reader is already familiar with this field, since only a concise high-level overview will be provided.
2.4.1 Basic principles
Without neglecting all the other important discoveries and breakthroughs, a fair statement of the main phenomenons responsible for the electromagnetism science foundation is that an electric current flowing through a conductor produces a magnetic field and that a changing magnetic field leads to a current flow in a closed-loop circuit linked by a magnetic field; this magnetic field can be decomposed into two vector quantities, the magnetic flux density B and the magnetic field intensity H.
The basic magnetic relationships will be presented through an analogy with electrical quanti-ties, as seen in figure 2.20.
The magnetomotive forceF between two points can be described as F =Z x2
x1
H · dl (2.5)
For a uniform magnetic field intensity H, previous equation becomes
F = H · l (2.6)
where l is the length of the element subject to the magnetic field. This is comparable to an uniform strength electric field E, which induces a voltage V = E · l between two points distant l from each other.
The total magnetic flux φ over a surface S with area Accan be deduced as
φ =
Z
sur f ace S
B · dA (2.7)
For a uniform magnetic flux density B, the previous equation becomes
φ = B · Ac (2.8)
In comparison with the electrical quantities, the magnetic flux density B can be seen as the electrical current density J, whereas the magnetic flux φ is comparable to the electric current I.
If one considers a coil with N turns in a magnetic field, accordingly to Faraday’s and Lenz law, the flux φ induces a voltage in the coil such as
v(t) = −Ndφ (t)
Figure 2.20: Analogy between magnetic and electrical quantities [21]
Ampere’s lawrelates the current in a closed contour with the magnetic field H as
I
C
H · dl = i (2.10)
where i is the total current enclosed by the contour.
A relationship between B and H can be established in respect to the core permeability µ, as
B = µH (2.11)
Core permeability µ is expressed in H/m and is usually expressed as the product of free air permeability µ0and the relative permeability µras
µ = µ0· µr (2.12)
where µ0= 4π × 10−7 H/m.
Figure 2.21 represents the B-H magnetisation curve of a typical magnetic core material. It can be verified that both saturation and hysteresis are present in this nonlinear characteristic. This behaviour is dependent of the used material and will be further explained in the next section.
2.4.2 Hysteresis loop and Core Materials
In a magnetic circuit, a magnetic core is used to conduct magnetic flux, whereas the relative permeability emerges as a factor of how much better the material is than a medium, such as air, for conducting magnetic flux. As stated in section 2.4.1, the relationship between the magnetic flux density B and the magnetic field intensity H is related to the core magnetic material.
The magnetisation curve in figure 2.21 represents the hysteresis loop in a ferromagnetic mate-rial; the main reason behind hysteresis is the "memory effect" inherent to ferromagnetic materials since they remain magnetised after the removal of the external magnetic field. An analysis of the
Figure 2.21: B-H characteristic curve - adapted from [22]
B-Hcurve can be initiated by increasing the magnetic field intensity H. By increasing H, the mag-netic flux density B will also increase until the saturation value Bsis reached, in which a complete
alignment of the magnetic moments is verified.
When reducing the magnetic field intensity H, a magnetic domains’ alignment can still be verified as a result of opposing forces. Therefore, the initial unaligned arrangement can only be obtained by further reducing the magnetic field intensity. When the magnetic field intensity H is reduced to zero, a residual magnetic flux density can be found in the core, whose magnitude Bris
known as residual flux density or remnant magnetisation. To return B to zero, a negative magnetic field intensity, −Hc, denominated coercive force must be applied. A saturation flux density value
−Bsis achieved if the magnetic field intensity keeps being decreased. A positive coercive force
+Hcis able to return the flux density to zero. As a result, the hysteresis loop is the obtained closed
loop as a result of the magnetic field intensity variation in a recurrent manner.
Two types of magnetic materials can be found in power electronic applications, soft magnetic materials and hard magnetic materials, whose characteristics and properties influence the shape of the hysteresis loop. In comparison with soft magnetic materials, hard magnetic materials show a wider hysteresis loop, higher values of coercive force Hc, thus making them primarily used in
permanent magnets applications. On the contrary, soft magnetic materials are employed in both power inductors and transformers cores due to the easy magnetisation and demagnetisation. Nar-rower hysteresis loop with inherent low values of coercive force can also be found. Before moving to an analysis of the different soft magnetic materials used in power electronics, the concept of Curie temperature must be provided; it is the temperature at which a ferromagnetic material starts to behave as paramagnetic, being an important parameter to take into account when designing a inductor/transformer. One can classify soft magnetic materials under five categories, which will be concisely present as follows [22, 23].
• Ferrites cores are widely used in power electronics applications, mainly due to their low cost. Low eddy-current losses are almost negligible due to high resistivity, usually 105Ω · m. Manganese-zinc (MnZn) and nickel-zinc (NiZn) are the two most common types of ferrite, being used in high-frequency applications up to 50 MHz. Nevertheless, drawbacks can be listed as low Curie temperature, ranging from 150 to 300oC whereas it can be expected low saturation flux density Bs, usually lower than 0.5 T.
• Powder cores are made by mixing iron or iron allow powder with an insulation material, to keep the particles separated from each other, thus resulting in a distribute air gap. As a result, low relative permeability is obtained, usually within the range of 3 to 550. Higher saturation flux density is verified as well as low eddy-current power losses. In comparison to ferrites, higher core losses and lower winding losses are expected.
• Nanocrystalline materials contain iron-based ultra-fine crystals, while combining high sat-uration flux density Bs with low losses at high frequencies. Usually relative permeability
is within the range 15000 to 150000, whereas saturation flux density Bsis between 1.2 and
1.5 T and Curie temperature occurs at 600oC. Nanocrystalline cores can be employed in applications up to 150 kHz.
• Laminated iron alloys are used in low to medium frequency applications. The most com-mon alloy is based on the addition of silicon for both resistivity increase and eddy-current losses reduction. Iron cores’ relative permeability is usually within 2500 to 5000, whereas the saturation flux density ranges from 1.5 to 2.2 T. Being used in applications up to 500 Hz, iron cores have their Curie temperature between 760oand 810oC.
• Amorphous alloys are also commonly known as metallic glasses due to their similar non-crystal structure. Relative permeability covers a range from 150000 to 1000000, whereas the saturation flux density is between 0.7 to 1.8 T. Maximum operating frequency is 250 kHz and low resistivity can be found to be around 1.2 µΩ · m.
2.4.3 Power losses
Magnetic devices’ losses can be categorised as winding or copper loss and core loss, which in-cludes hysteresis loss and eddy current loss. Starting by copper loss, at low frequencies, heat is dissipated in a wire used to build a winding as
Pcu= R · I2 (2.13)
where I is the RMS current and R the wire resistance. However, at high frequencies, current in a conductor is not uniformly distributed, instead it tends to concentrate near the conductor surface. This is the so called skin effect and emerges inside a conductor to oppose the AC flux in the form of eddy currents. As a result, for very high frequencies, only a thin layer of conductor, skin depth, is conducting current, which increases the necessary conductor effective resistance.
On a different note, operating at high frequencies also result in the proximity effect, in which the time-varying current in one conductor interferes with the current distribution of nearby con-ductors, thus increasing even further the overall resistance. A countermeasure for both skin and proximity effect is to employ insulated stranded wire, such as Litz wire.
Regarding core loss, hysteresis loss is due to the remaining energy used to reorientate the mag-netic domains of a magmag-netic material; thereby, hysteresis loss is proportional to the B-H hysteresis area, core volume and frequency. Soft magnetic materials usually present narrower hysteresis loops than hard magnetic materials, thus resulting in lower hysteresis loss at a given frequency. At last, eddy current loss surge due to induced currents circulating in the core; they are proportional to the square of frequency, and directly proportional to the cross-section area of the core. Insulated laminations are often employed for reducing eddy current loss. Total core loss, including both hysteresis and eddy current loss, can be described by the Steinmetz equation as
Pf e= KcfαBβmax (2.14)
where Pf e is the core loss per unit volume, Bmax is the peak value of the flux density under
sinusoidal excitation at frequency f and Kc, α and β are parameters provided by the manufacturer.
2.4.4 Transformer modelling
Numerous applications for a transformer regarding power electronics applications can be listed. Bearing in mind the focus of this dissertation, main features rely on energy conversion and con-trol, electrical isolation between parts operating at different potentials within a system as well as providing multiple outputs. Furthermore, a general rule of thumb is that transformer’s overall size is inversely proportional to its operating frequency [22].
A two-winding transformer is represented in figure 2.22, consisting of two coils wound around a common magnetic core. Coil 1 is made of N1turns and coil 2 is made of N2turns, being referred
as primary and secondary winding respectively.
Figure 2.22: Two-winding transformer [23]
A first analysis will be carried out by considering that an ideal transformer is lossless, in which no magnetic energy is stored, core relative permeability is infinite, all losses are assumed
